Criado por Olli Niemitalo em 2003-01-21, última modificação 2012-08-04 Em 1998, eu tive algum tempo extra, enquanto outros estavam lendo para exames finais do ensino médio, e entrou em processamento de sinal digital. Eu escrevi como aprendi, e aqui está o resultado. Não é inteiramente preciso em lugares, mas pode servir como um bom tutorial para o mundo do áudio DSP. Anteriormente este documento foi chamado Yehars digital som processamento tutorial para o braindead, mas eu tenho um pouco cresceu fora da minha identidade cena ao longo dos anos. Aprecie a arte ASCII Isto é escrito para os entusiastas de processamento de sinal de áudio digital (como o título sugere) e outros que precisam de informações práticas sobre o assunto. Se você não tem isso como uma experiência de leitura linear e encontrar dificuldades, verifique se há algo para ajudá-lo nos capítulos anteriores. Nos gráficos de frequência de filtro, são utilizadas escalas de frequência e magnitude lineares. As alterações de página foram criadas para 60 impressoras de páginas de linha. Capítulo Shuffling IIR equações é escrito por meu irmão mais velho Kalle. E, graças a Timo Tossavainen por compartilhar seu conhecimento de DSP Copie e use este texto livremente. Por Olli Niemitalo, oiki. fi Observe que a amostra pode significar (1) um som amostrado ou (2) um ponto de amostra Dados de som amostrados é uma pilha de amostras, valores de amplitude tomados da onda sonora real. A taxa de amostragem é a freqüência dos disparos. Por exemplo, se a frequência for 44100, foram tomadas 44100 amostras num segundo. Heres um exemplo de amostragem: O som original é a curva, e 0s são os pontos amostrados. A linha reta horizontal é o nível zero. Um som amostrado só pode representar freqüências de até metade do samplerate. Isso é chamado de freqüência de Nyquist. Uma prova simples: Você precisa ter armazenado pelo menos dois pontos de amostra por ciclo de onda, a parte superior ea parte inferior da onda para poder reconstruí-la mais tarde: Se você tentar incluir as freqüências Nyquist acima em seu som amostrado, tudo o que você obtém É distorção extra como eles aparecem como freqüências mais baixas. Um som consiste de componentes de frequência. Todos eles parecem exatamente como ondas senoidais, mas eles têm freqüências, fases e amplitudes diferentes. Vamos olhar para uma única freqüência: Agora, tomamos a mesma freqüência de outro som e observamos que ele tem a mesma amplitude, mas a fase oposta (girada 180 graus). A fusão de dois sinais é feita simplesmente adicionando-os juntos. Se fizermos o mesmo com estas duas ondas senoidais, o resultado será: Ele fica em silêncio. Se pensarmos em outros casos, onde a diferença de fase é menor que 180 graus, obtemos ondas senoidais que têm diferentes amplitudes e fases, mas a mesma freqüência. Aqui está a maneira de calcular a fase e a amplitude da onda senoidal resultante. Converta a amplitude e a fase em um número complexo, onde o ângulo é a fase e o valor absoluto a amplitude. Se você fizer isso com ambas as ondas senoidais, poderá adicioná-las como números complexos. Como você vê, a fase da nova onda senoidal é de 45 graus e a amplitude sqrt (1212) sqrt (2) cerca de 1,4. É muito importante que você entenda isso, porque em muitos casos, é mais prático apresentar a amplitude e A fase de uma frequência como um número complexo. Ao adicionar dois sons amostrados juntos, você pode realmente apagar algumas freqüências, aquelas que tiveram fases opostas e amplitudes iguais. A amplitude média do som resultante é (para originais independentes) sqrt (a2b2) onde aeb são as amplitudes dos sinais originais. O uso principal de um filtro é escalar as amplitudes dos componentes de freqüência em um som. Por exemplo, um filtro de passagem baixa silencia todos os componentes de freqüência acima da freqüência de corte, ou seja, multiplica as amplitudes por 0. Ele permite que todas as freqüências abaixo da freqüência de corte não sejam atenuadas. Se você investiga o comportamento de um filtro de passagem baixa, movendo várias ondas de diferentes freqüências através dela, e medir as amplificações, você obtém a resposta de freqüência de magnitude. Aqui está um gráfico da curva de resposta em frequência de magnitude de um filtro passa-baixa: A frequência está no eixo e a amplificação no eixo. Como você vê, a amplificação (escala) das freqüências abaixo da freqüência de corte é 1. Assim, suas amplitudes não são afetadas de forma alguma. Mas as amplitudes de freqüências acima da freqüência de corte são multiplicadas por zero, então elas desaparecem. Os filtros nunca adicionam novos componentes de freqüência ao som. Eles só podem escalar as amplitudes de freqüências já existentes. Por exemplo, se você tem uma amostra completamente silenciosa, você não pode obter qualquer som dele por filtragem. Além disso, se você tiver uma amostra de onda senoidal e filtrá-lo, o resultado ainda será a mesma onda senoidal, apenas talvez com diferentes amplitude e fase - nenhuma outra freqüência pode aparecer. Profissionais nunca se cansam de lembrar-nos como é importante não esquecer a fase. Os componentes de freqüência em um som têm suas amplitudes e. Fases. Se tomarmos uma onda senoidal e uma onda coseno, vemos que eles se parecem, mas eles têm uma diferença de fase de pi2, um quarto de um ciclo completo. Além disso, quando você tocá-los, eles soam iguais. Mas, tente usar um fone de ouvido e tocar a onda senoidal no canal esquerdo ea onda de coseno no canal direito. Agora você ouve a diferença Fase em si não contém informações importantes para nós assim que não é ouvido, mas a diferença de fase, de uma freqüência, entre as duas orelhas pode ser usado na estimativa da posição da origem do som para que seja ouvido. Filtros têm uma resposta de freqüência de magnitude, mas eles também têm uma resposta de freqüência de fase. Heres uma curva de exemplo que poderia ser de um filtro de passagem baixa: Se você filtrar um som, os valores da resposta de freqüência de fase são adicionados às fases das freqüências do som original. Linear (linha reta) fase é a mesma coisa que um simples atraso, embora possa parecer selvagem na trama, se ele vai em torno de várias vezes. Se o seu filtro de passagem, por exemplo, não tem uma resposta de freqüência de fase linear, você não pode transformá-lo em um filtro highpass simplesmente subtraindo sua saída do original com o mesmo atraso. Matemática complexa com filtros A resposta de um filtro para uma única frequência pode ser expressa como um número complexo, onde o ângulo é a resposta de fase do filtro eo valor absoluto da resposta de magnitude. Quando você aplica o filtro a um som, você realmente faz uma multiplicação complexa de todos os componentes de freqüência no som pelos correspondentes valores de resposta de filtro. (Leia o capítulo Adicionando duas ondas sinusoidais juntas se achar que isso é difícil de entender.) Exemplo: A resposta de um filtro é (0,1) a 1000Hz. Você filtra uma onda senoidal, com as informações de amplitude de amplificação de fase apresentadas como o número complexo (0,1), da mesma freqüência com ele: A fase da onda senoidal foi girada 90 graus. Sem alteração na amplitude. Combinando filtros A resposta combinada destes dois filtros colocados em série é a resposta de A multiplicada pela resposta de B (números Complexos como sempre). Se você só precisa saber a resposta de magnitude, você também poderia multiplicar os valores absolutos. Na figura, ambos os filtros obtêm suas entradas da mesma fonte. Suas saídas são então adicionadas novamente juntas, formando a saída final. Agora você precisa usar adição para resolver a resposta combinada. O filtro FIR é mais direto e mais fácil de entender. A resposta ao impulso finito significa que quando a entrada do filtro permanece zero durante um certo tempo, a saída do filtro também se torna zero. Um filtro de resposta de impulso infinito nunca se estabelece completamente depois de desligar a entrada, mas ele fica mais silencioso e silencioso. Um filtro básico de FIR poderia ser: onde input significa os valores de amostra alimentados para o filtro. Neste caso, as pessoas falam de um filtro de 3 tap. Seu até os coeficientes (a0, a1, a2) o que este filtro fará ao som. Escolher os valores dos coeficientes é a parte mais difícil, e bem chegar a isso mais tarde. Para projetar seus próprios filtros, você precisa entender algumas das matemáticas por trás e conhecer os métodos certos. No exemplo de filtro acima, apenas os valores de entrada passados são usados. Em filtros em tempo real, este é um requisito, porque você não sabe as entradas futuras. Em editores de amostra e tal, você não tem essa limitação, porque você tem todos os dados de entrada pronta quando você começar. Se o seu filtro é: e você precisa de uma versão em tempo real do mesmo, basta convertê-lo para: A única diferença é o atraso de uma amostra no filtro em tempo real. Ao contrário dos filtros FIR, os filtros IIR também usam seus valores de saída anteriores ao criar sua saída atual. Heres um exemplo simples: Isso poderia ser chamado de 3 entrada, filtro de tap de 3 saídas. Os filtros IIR nunca podem usar valores de saída futuros, porque esses ainda não existem. Pode haver várias maneiras de implementar o mesmo filtro IIR. Alguns podem ser mais rápidos do que a maneira usual de entrada-saída-e-coeficientes. De qualquer forma, cada filtro IIR pode ser escrito neste formulário, e deve ser usado no projeto do filtro e exames de cálculos. Uma resposta de impulso (O que o filtro fará a um impulso de um ponto de amostra) de um filtro IIR muitas vezes parece mais ou menos assim nos dados de amostra: Alguns filtros IIR mal projetados são instáveis. Isso resulta em ouput ficando mais alto e mais alto em vez de mais silencioso e mais silencioso. Um exemplo simples disso é: output (t) input (t) 2output (t-1). Assim que recebe dados de entrada, fica louco. Os tipos de filtro acima descritos processam a amostra de dados por amostra. Não é assim, se você implementar seu filtro usando FFT, Fast Fourier Transformation. FFT geralmente opera em pedaços de comprimento 2n. Primeiro, você deve ter sua resposta de impulso de filtro planejada pronta. Em seguida, converta-o, usando FFT, para informações espectrais - números complexos representando as fases e amplitudes dos componentes de freqüência. Esses componentes são chamados bins, porque suas freqüências são fixas e uniformemente distribuídas, e se os dados originais continham quaisquer freqüências intermediárias, então a maior parte da energia dessa freqüência será distribuída entre as caixas próximas. Agora, você FFT também os dados de amostra que você deseja filtrar, e multiplicar os intervalos de freqüência resultante com aqueles do filtro. Então IFFT (Inverse FFT) é usado para converter as informações em um pedaço de dados de amostra filtrada. Assim, a multiplicação dos dois dados do domínio da frequência resultou na convolução dos dois dados do domínio do tempo. No entanto, há uma captura: FFT opera em sinais periódicos, ou seja, se você tiver uma resposta de impulso do filtro, enquanto o FFT pedaço, em seguida, qualquer não-zero dados amostra no meio do FFT pedaço resultará com a convolução envolvendo a cauda Do filtro em torno do limite FFT. Para evitar este problema, você pode usar FFT duas vezes o tempo que a resposta de impulso do filtro, e ao fazer FFT nos dados da amostra, apenas preencher o buffer de entrada FFT a meio caminho e definir restante da entrada para zero. Para entradas mais longas, você processaria os dados em pedaços como esse e adicionaria os pedaços filtrados resultantes juntos. Isso é chamado de método de sobreposição-adição. Outra opção é sobreposição-salvar (procure-o se você gosta). FFT também pode ser usado para analisar o conteúdo de freqüência de dados de amostra, por qualquer motivo. Se você tomar apenas um pedaço de dados de amostra, ele tem bordas afiadas, o que é ruim para FFT. As funções Windowing são usadas para suavizar estas arestas. Coseno elevado, cos (x pi2) 2, é uma função de janela possível. Aqui você vê o que acontece quando você aplica essa função de janela para um pedaço de dados de amostra: Às vezes (reamostragem, precisamente definido atraso) você precisa para obter amostras de valores entre os samplepoints conhecidos. Isso é quando você precisa de interpolação. Se você não interpolar, e apenas jogar fora a parte fracionária de seu sampleoffset, você começa um monte de distorção de alta freqüência: No exemplo, os samplepoints originais tentam representar uma onda senoidal. Quanto mais próxima a curva interpolada for a uma onda seno, melhor será o algoritmo de interpolação. O método de interpolação mais simples é a interpolação linear. Linhas retas são desenhadas entre dois pontos de amostra adjacentes: Ainda parece bastante nervoso para ser uma onda senoidal. No entanto, a melhoria para não-interpolada é significativa. Há também uma desvantagem - as freqüências logo abaixo da freqüência de Nyquist obter atenuado, ainda mais do que sem interpolação. Aqui está a fórmula para a interpolação linear: new (int) (old (int1) - old (int)) fract, onde int significa a parte inteira do offset da amostra e fract a parte fracionária. O próximo passo poderia ser a curva de Hermite, que dá em todos os sentidos melhor qualidade do que a interpolação linear: Com a interpolação linear, você precisava saber 2 samplepoints a tempo de ser capaz de desenhar a linha. Com a curva de Hermite, o número é 4. A curva de interpolação passa pelos dois pontos médios, e os pontos 1 e 4 são usados na formação da curva. A fórmula é um cubic: E este aqui é onde a, b, c, d foram resolvidos de: Uma interpolação perfeita também existe. Ao substituir todos os pontos de amostra com curvas de sinc corretamente escalonadas, sin (pi x) (pi x), e adicionando-os juntos, você obtém a interpolação exata e perfeita. Aqui está um dos samplepoints substituídos por uma curva escalada de sinc: A curva de Sinc é infinitamente longa, então você terá que usar todos os samplepoints no cálculo de um valor interpolado. Uma solução prática seria limitar o número de amostras para dizer 1000. Ainda será muito lento para uma aplicação em tempo real, mas itll dar grande precisão. Se você insiste em usar sinc em um algoritmo de interpolação em tempo real, tente usar uma função de janela e um número baixo (pelo menos 6) de curvas de sinc. Downsampling Se você quiser downsample (diminuir o samplerate), você deve primeiro filtrar afastado as freqüências Nyquist acima, ou eles aparecerão como distorção na amostra downsampled. No processo de design do filtro, muitas vezes você precisa fazer compromissos. Para ter bordas afiadas ou inclinações íngremes na resposta de magnitude, você precisará de um filtro grande e, portanto, lento. Em outras palavras, filtros com baixo número de torneiras praticamente sempre têm respostas de magnitude suavemente inclinadas. No caso de filtros IIR, as arestas em grandeza geralmente significam uma resposta de frequência de fase feia (muito não linear) e uma resposta de fase próxima a linear uma resposta de magnitude suavemente inclinada. Com filtros FIR, uma tentativa de criar arestas muito afiadas pode causar acenando na magnitude das frequências próximas. Os filtros IIR são ótimos para uma rotina em tempo real, porque são rápidos, suas propriedades (por exemplo, freqüência de corte) podem ser rapidamente alteradas no meio da ação e soam como verdadeiros filtros analógicos. ) A resposta de fase não linear de filtros IIR geralmente não importa. Os filtros FIR podem ser usados onde a qualidade ea fase linear são importantes, por exemplo, em um editor de amostras. As pessoas que filtram outros sinais do que o som, muitas vezes desejam resposta de freqüência de fase linear. Com sinal estéreo, é importante ter alterações de fase idênticas nos canais esquerdo e direito. Alguns filtros e suas respostas estilizadas de freqüência de magnitude: Se você tem um programa de cálculo simbólico, eu recomendo fortemente que você usá-lo nos cálculos mecânicos, apenas para tornar sua vida mais fácil. Derive é um antigo programa DOS, mas ainda é muito útil. Ruído branco O ruído branco significa o tipo de ruído que tem espectro plano. Você pode facilmente criá-lo usando números aleatórios como valores de amostra. Se você quiser saber a resposta de freqüência de magnitude de um filtro, aplicá-lo em uma longa amostra de ruído branco e, em seguida, executar uma análise de espectro na saída. O que você vê é a resposta de freqüência de magnitude do filtro. Outra maneira é enviar um impulso de uma amostra, que originalmente tem um espectro plano. Um impulso se parece com isto nos dados de amostra: 0, 0, 0, 0, 1, 0, 0, 0, 0 - onde o impulso é o 1 no meio. A partir dos dois, o impulso thingy é mais rápido, mas usando ruído branco pode dar resultados mais limpos, porque os erros serão menos visíveis. Por muito as mesmas razões, quando você está assistindo vídeos, uma imagem estática será mais nevado do que a imagem em execução. Fazer uma análise de espectro em uma amostra longa geralmente é feito dividindo-o em pedaços menores, analisando-os separadamente e depois tomando a média de todas as análises. Minha escolha pessoal aqui seria o programa Cool Edit 96, que é para o Windows. O método Pole-zero é a maneira mais fácil de projetar filtros IIR rápidos e simples. Quando você tiver aprendido, você será capaz de projetar filtros por si mesmo. Aqui está o plano Z complexo, o usado no método pólo-zero: Imagine as freqüências a serem enroladas ao redor do círculo unitário. No ângulo 0 temos 0Hz, em pi2 temos samplerate4, em pi temos samplerate2, a frequência de Nyquist. Você não deve se preocupar com freqüências mais altas, uma vez que nunca aparecerá no sinal, mas de qualquer maneira, em 2pi (ciclo completo), temos a freqüência de amostragem. Portanto, se você usou a freqüência de amostragem 44100 Hz, 0 Hz estaria em (1,0), 11025 Hz em (0,1) e 22050 Hz em (-1,0). O que são pólos e zeros então Eles são pequenas coisas bonitas que você pode colocar no Z-plano, assim: Existem algumas regras que você tem que lembrar. Os poloneses devem estar sempre dentro do círculo da unidade, nunca fora ou sobre ele. Os zeros podem ser colocados em qualquer lugar. Você pode usar qualquer número de pólos e zeros, mas todos devem ter pares conjugados, se não estiverem posicionados no eixo. Conjugado pares significa que se você colocar por exemplo um zero para (0,6, 0,3), você deve colocar outro zero para a coordenada conjugada, (0,6, -0,3). E a mesma coisa com pólos. Mas hey O que fazem pólos e zeros DO Poles amplificam freqüências, zeros atenuam. Quanto mais próximo um pólo estiver a uma freqüência (no círculo da unidade, lembre-se), mais ele se amplifica. Quanto mais próximo um zero é a uma freqüência, mais ele fica atenuado. Um zero no círculo da unidade silencia completamente a freqüência na qual ele está sentado. Agora pode ser o momento certo para tentar isso sozinho. Há programas de design de filtro livre em torno de que permitem que você jogue com pólos e zeros. Um candidato poderia ser: Demo QEDesign 1000 para Windows. Seu em algum lugar no Internet, youll o encontra. Projetando um filtro passa-faixa O filtro mais simples projetado usando pólo-zero é o seguinte filtro passa banda: Os polos amplificam as freqüências, assim você poderia tirar a conclusão de que a freqüência mais amplificada é a do mesmo ângulo que o pólo. E você está quase certo O único problema vem do pólo conjugado, que também dá a sua própria amplificação. O efeito é mais forte em ângulos próximos de 0 e pi, onde a distância entre os dois pólos é a menor. Mas não deixe isso confundi-lo, bem voltar para ele mais tarde. Assim, o ângulo do pólo determina a freqüência de banda passante. Qual é o efeito do valor absoluto (raio) então Como dito, os pólos amplificam as freqüências ea amplificação é mais forte quando o pólo está mais próximo de uma freqüência. Em nosso filtro passa-banda, aumentar o raio do pólo faz com que a resposta de magnitude se torne mais íngreme e passband mais estreita, como você vê aqui: Posições de pólos: Correspondente magnitude parcelas resposta de freqüência (normalizado): Vamos chamar o raio r de agora em diante. (Alguns de vocês podem se lembrar da letra q de filtros analógicos, ressonantes, o que é praticamente o mesmo). Neste caso temos a limitação: 0 r lt 1, uma vez que os pólos devem estar dentro do círculo da unidade. Então mudar r muda inclinação, ressonância. Esta ressonância - não é uma coisa mágica, apenas uma freqüência sendo amplificada mais do que outros. De pólos e zeros a coeficientes de filtro Existe uma função de transferência: onde z é freqüência, na forma coordenada (complexa) envolvida em torno da unidade-círculo. H (z) dá a resposta (complexo) do filtro na frequência z. P1, p2, p3 e assim por diante são posições de pólos e z1, z2, z3 e assim por diante posições de zeros. A0 é o primeiro coeficiente de entrada do filtro. Aqui está a fórmula do filtro IIR, caso você tenha esquecido: Nosso filtro passa-banda tem apenas um pólo e seu par conjugado, então podemos simplificar a função de transferência e substituir p1 e p2 pelas coordenadas dos pólos conjugados: Divisor um olhar mais atento. Digamos: Poderes de z aqui são realmente índices para a saída do filtro: Então sabemos calcular os coeficientes do lado de saída a partir da posição do pólo: OK Vamos dizer que a frequência de passagem está no plano Z na posição ph: O pólo está no mesmo ângulo que a freqüência no círculo da unidade, mas tem o raio r. Portanto: Agora que sabemos como a posição do pólo depende da freqüência, podemos reescrever os coeficientes do lado da saída: Mas não devemos esquecer o dividendo (da função de transferência), onde as potências de z são índices para a entrada do filtro : Isso deve ser adicionado ao que já resolvido do lado da saída: Em seguida, temos que decidir o que colocar para a0. Isso é chamado de normalização. O objetivo de a0 é apenas dimensionar a saída do filtro. Em nosso filtro passa-faixa, queremos que a amplificação na freqüência de banda passante seja 1. Então podemos escrever a equação: Lá está Agora o filtro está pronto: Melhorando o filtro passa-banda simples Poderíamos compensar o efeito do pólo conjugado adicionando um Zero no eixo, entre os pólos. Por exemplo, se tivéssemos pólos nas coordenadas (0.6, 0.5) e (0.6, -0.5), colocar um zero em (0.6, 0): A função de transferência para isto é: Os coeficientes do lado de saída são exatamente os mesmos que antes . Os coeficientes do lado de entrada podem ser resolvidos da seguinte forma: No caso de você querer usar este filtro, você deve ser capaz de fazer a normalização você mesmo. Eu não vou fazer isso aqui. Palavras de sabedoria É fácil fazer um filtro mais eficiente: Dobro todos os pólos e zeros. A resposta de freqüência do novo filtro é o quadrado do antigo. Existem maneiras melhores, mas este é o mais fácil. Se você colocar um zero em um poste, você neutraliza ambos. Um pólo fora do círculo da unidade faz com que o filtro se torne instável. Um pólo no círculo da unidade pode transformar o filtro em um oscilador. Grande número de pólos e zeros significa grande número de torneiras. Zeros afetam os coeficientes de entrada, saída de pólos. Pólos e zeros devem ter pares conjugados, porque caso contrário você terá coeficientes de filtro complexos e, conseqüentemente, sinal de saída complexo. Com baixos valores de r, a freqüência mais amplificada nem sempre está no mesmo ângulo com o pólo, devido ao efeito do pólo conjugado. Tente diferenciar a resposta de magnitude se você deseja precisão exata. Um filtro IIR sem pólos é um filtro FIR. 0 r lt 1 sempre se aplica. Bandpass com r Leia o capítulo IIR usando o método de pólo-zero. Notch com r Quanto maior o r, mais estreita é a stopband. Lowpass com r Isso pode ser feito de várias maneiras: Quanto maior o r, mais forte a ressonância. Resonant lowpass filtro é certamente o tipo de filtro mais utilizado em sintetizadores. Se você ler sobre interpolação de sinc no capítulo Interpolação de som amostrado, você sabe que você pode substituir um único pico de amostra (impulso) nos dados de amostra com uma função sinc corretamente esticada. Corretamente alongado significa amplitudesinc (t). Quando você executa uma análise de espectro em um impulso, você obtém um espectro plano com limite superior em samplerate2, a freqüência de Nyquist. Porque impulso sinc, este é também o espectro de sinc: Você poderia tirar a conclusão de que você começa a função sinc se você somar todas as freqüências de 0 a SR2, e dividir a soma pelo número de freqüências, para cumprir a equ (0) 1. E você vai estar certo. A partir da análise de espectro, você sabe que todas as freqüências somadas juntas têm amplitudes iguais. Mas qual é a sua fase no centro do impulso Sinc função é simétrica em torno de x0, por isso é coseno - assim sinc deve ser feito de cossenos. Se você testar isto com cerca de 100 cossenos, obtém uma aproximação bastante próxima de sinc. A soma de todas as freqüências de 0 a 1 (comparável a SR2), dividida com seu número, pode ser escrita como: (Aqui oo significa infinito) Como feito acima, x deve ser substituído por pi t, porque o comprimento do ciclo do pecado é 2 pi, que deve ser esticada para 2 (que é o comprimento de onda da frequência de Nyquist nos dados da amostra). Fase de deslocamento E se substituíssemos os cossenos por senos Vamos tentar Há uma fórmula universal (que, por exemplo, eu me inventei), podemos usar: Agora, se substituirmos todos os impulsos do som por esta nova função, Deslocamento de fase de -90 graus Isso pode ser feito criando um filtro FIR, onde os coeficientes são tomados dessa nova função: (1-cos (pi t)) (pi t), mas em ordem inversa, substituindo t por - t , Então ele se torna: (cos (pi t) -1) (pi t). Heres um exemplo que explica por que é necessário usar - t em vez de t: Digamos que você deseja substituir todos os impulsos no sinal com a seqüência 1,2,3. Se o sinal de entrada é 0,1,0, o senso comum diz que deve se tornar 1,2,3. Se você apenas usa 1,2,3 como coeficientes de filtro nessa ordem, o sinal filtrado torna-se: O que não é o que você pediu Mas se você usar coeficientes 3,2,1, você obtém o resultado correto, Ok, de volta ao Filtro de mudança de fase de 90 graus. Quando você está escolhendo os coeficientes de filtro de (cos (pi t) -1) (pi t), em t0 você infelizmente obter uma divisão por zero. Evite isso calculando o limite t-gt0, no papel, ou com um proggy de matemática. Se você usa seu cérebro um pouco, você percebe que é 0, porque a fórmula do filtro é uma soma de senos, e sin (0) 0, então em t0 é uma soma de zeros. Como sinc, nossa nova função não tem fim, então um compromisso deve ser feito no número de torneiras. Isso causa ondas na resposta de magnitude e atenuação das freqüências mais baixas e mais altas. Aplicando uma função windowing aos coeficientes, você pode começar livrado das ondas, mas eu não sei qualquer coisa que ajudaria com a atenuação, exceto mais torneiras. As funções de janelas usadas com o FFT funcionam aqui também. O centro da função windowing deve estar em t0, e ele deve ser esticado de modo que as arestas sejam colocadas no primeiro e no último toque. Você também pode obter um desvio de fase de qualquer ângulo a: Note que a inversão de t já foi feita aqui, então podemos pegar os coeficientes diretamente dessa fórmula. O limite t-gt0 é naturalmente cos (a), porque todos os cossenos somados tiveram fase a em x0. No caso de você ainda não perceber, a idéia principal na filtragem de FIR é criar uma função que contém as freqüências que você deseja passar a filtragem. As amplitudes das freqüências na função definem diretamente a resposta de freqüência de magnitude do filtro. As fases das freqüências definem a resposta de fase. A inversão dos coeficientes só é necessária com filtros de mudança de fase, porque filtros que não introduzem um deslocamento de fase de qualquer tipo são simétricos em torno de t0. Definindo a faixa de freqüência incluída Se você usar sinc como sua fórmula de coeficiente de filtro, você realmente não faz nenhuma filtragem, porque todas as freqüências de 0 a Nyquist são igualmente apresentadas em sinc. Aqui você verá como você pode selecionar quais freqüências estarão presentes em sua fórmula de coeficiente de filtro. Lembre-se de onde originalmente obtivemos sinc de: Na integral, o limite superior (1x) realmente representa a freqüência mais alta incluída (1), eo limite inferior (0x) o menor (0). Então, se você quer uma fórmula para um filtro passa-banda, você pode escrever: onde top e bottom são as freqüências de corte de tal forma que 1 significa a freqüência de Nyquist e 0 significa 0Hz. Agora, basta colocar as frequências que quiser, calcular e substituir x com (pi t). Por exemplo, se você quiser fazer um filtro passa-baixa de halfband (que naturalmente tem frequência de corte no samplerate4, o mesmo que a freqüência de Nyquist 2): Para criar filtros multi-band, você pode combinar várias fórmulas de filtro passa banda adicionando Juntos. O exemplo do equalizador Se você quiser fazer um equalizador (um filtro que permite definir as magnitudes para certas freqüências), você provavelmente juntará um monte de fórmulas de filtro passa-banda, dimensionadas pelas magnitudes que você deseja para os segmentos de freqüência. Isso lhe dá uma resposta de magnitude que se parece muito com como se fosse feito de tijolos: Talvez você gostaria que ele se pareça mais com isto em vez disso: Existem três maneiras. A primeira maneira é usar tijolos menores, o que significa que você divide a freqüência em segmentos mais estreitos do que antes e usar a interpolação para obter os valores de magnitude para os novos filtros de banda estreita que você combina. A segunda maneira é definir um polinômio (como ax3bx2cxd) que tem as características desejadas (e onde x1 representa freqSR2), e fazer a resposta de magnitude do seu filtro para segui-lo. Isso é possível. A terceira maneira é juntar várias fórmulas de filtros de rampa de banda. Na resposta de magnitude esta solução parece linhas retas desenhadas entre as freqüências definidas adjacentes. Isso também é possível, e, na minha opinião, a melhor solução. Em sinc, todas as ondas de coseno juntas têm amplitudes iguais, como você vê aqui - todas as freqüências são tratadas igualmente: Você pode mudar isso colocando uma função g () que define as amplitudes das ondas de coseno De frequências diferentes: Se a função g (x) for forma axb, os cálculos vão assim: Para um exemplo simples, se queremos que a resposta de freqüência de magnitude seja uma reta, começando de 0 a 0Hz e terminando em 1 em SR2, definimos g (x) x: E os cálculos de fórmula de coeficiente de filtro para isso: Em outros casos, para obter a fórmula para um polinômio completo, faça os cálculos para cada um de seus termos (axb) separadamente e somar os resultados. Amplitude-faixa de passagem de faixa Heres um exemplo da resposta de freqüência da magnitude de um filtro da rampa: Para fazer uma rampa de passagem de faixa, você deve primeiramente definir o polynomial g (x) que descreve como a magnitude se comporta dentro dos limites do bandpass. A magnitude é linear dentro dos limites, então o polinômio g (x) deve ser a forma cxd. C e d podem ser resolvidos a partir das equações: onde x1 é o limite de frequência inferior, e x2 o mais elevado. Y1 e y2 são as magnitudes das frequências limite. Lembre-se que aqui x1 é igual a freqüênciaSR2. OK, aqui estão c e d resolvidos: G (x) cxd é um polinômio, e você já sabe como fazer a resposta de freqüência de magnitude tem a mesma forma (seção polinomial em forma de resposta de freqüência de magnitude) como um polinômio. Você também já sabe como incluir apenas um intervalo limitado de freqüências (Seção Definindo a faixa de freqüência incluída) em sua fórmula de coeficientes. Combine esse conhecimento, e você pode escrever a fórmula de coeficiente para o filtro de faixa de rampa: Uma nota sobre a implementação do equalizador. Se o equalizador for ajustável em tempo real, recalcular toda a fórmula do filtro do equalizador com todas as funções trigonométricas pode resultar muito pesado. Pode ser melhor precalcular os coeficientes para vários filtros sobrepostos, por exemplo, estes para um equalizador de três canais: Ao calcular os coeficientes para todo o equalizador, basta escolher os coeficientes correspondentes a partir destes, a escala de acordo com os controles deslizantes do equalizador e soma. Se você tomar seus coeficientes de filtro FIR diretamente de sua fórmula de filtro, você obtém uma resposta de magnitude muito ondulada. A razão é simples: o número de coeficientes é limitado, mas a fórmula do filtro não é, ele continua tendo valores diferentes de zero fora do intervalo que você está usando para os coeficientes. Uma função windowing ajuda. Not using a windowing function is the same thing as using a rectangular ( flat inside its limits) windowing function. Using a windowing function means that you multiply the values taken from your infinitely long filter formula by the corresponding values taken from your finitely long windowing function, and use the results as filter coefficients. Here are some windowing functions, and the produced magnitude responses of a FIR lowpass filter with a low number of taps, illustrated: As you see, the steeper the cutoff, the more waves you get. Also, if wed look at the magnitude responses in dB scale, wed notice that from the three, cos4 gives the best stopband ( the frequency range that should have 0 magnitude) attenuation. Mathematically, multiplication in the time domain is convolution in the frequency domain, and windowing is exactly that. (Also, multiplication in the frequency domain is convolution in the time domain.) I hope i didnt slam too many new words to your face. Time domain means the familiar time-amplitude world, where we do all the FIR and IIR filtering. The frequency domain means the frequency-amplitudeampphase world that you get into through Fourier transformation. And convolution In the time domain, FIR filtering is convolution of the input signal with the filter coefficients. Say you convolute 0,1,0,0,2,0,1,0 with 1,2,3 (where 2 is at the center): Youll get 1,2,3,2,4,7,2,3. If you understand this example, you surely understand convolution too. Ideally (impossible), there would be no windowing, just the constant value 1 infinitely in time. And a steady constant value in the time domain is same as 0Hz in the frequency domain, and if you (in the frequency domain) convolute with 0Hz, it is the same as no convolution. Convolution in the frequency domain equals to multiplication in the time domain, and convolution in the time domain equals to multiplication in the frequency domain. Sounds simple, eh But note that in this frequency domain, there are positive AND NEGATIVE frequencies. Youll learn about those in chapter Positive and negative frequencies. Words of wisdom You get flat (but not necessarily continuous) phase response if your filter (filter coefficients) is symmetrical or antisymmetrical (sides are symmetrical but have opposite signs, and the center crosses zero) around t0, even if you limit the number of coefs andor window them. Sometimes you can optimize your filter code a lot. Some coefficients may turn zero, so you can skip their multiplications. If your filter is symmetrical around t0, you can instead of input(t)ainput(-t)a write (input(t)input(-t))a). If your filter is antisymmetrical around t0, replace input(t)a-input(-t)a) with (input(t)-input(-t))a. Sinc(t) is 1 at t0, and 0 at other integer t values. Calculating the limit t-gt0 is very simple. If your filter formula was originally a sum cosines (meaning its not a phase shift filter), the limit t-gt0 is simply the area of the magnitude frequency response, in such way that the area of no filtering is 1. The actual filter implementation (after possible coefficient calculations) depends much on how the input data is fed to the filter. I can see three cases: You have the whole input data in front of you right when you start. A sample editor is a good example on this. This is the easiest case. With FIR filters, just take values from the input data, multiply with coefficients and sum, like this: output(t) a0input(t-2) a1input(t-1) a2input(t) a3input(t1) a4input(t2). The only problem is what to do at the start and at the end of the input table, because reading data from outside it would only cause problems and mispredictability. A lazy but well working solution is to pad the input data with zeroes, like this: This is how its mostly done with FFT filtering. With FIR filters, it isnt that hard to write a version of the routine that only uses a limited range of its taps, like this: and to use that version at the start and at the end. For this, it is easiest if you have a table of coefficients instead of hard-coding them into the routine. Data is fed to the filter in small chunks, but it is continuous over the chunk borders. This is the most common situation in programs handling realtime audio. One sample at a time. Case 2 can be treated as this, because the chunks can always be chopped into single samples. It is a fact that you cannot use future inputs in this case, so a FIR filter would have to be of form such as: output(t) a0input(t-4) a1input(t-3) a2input(t-2) a3input(t-1) a4input(t). Clearly this kind of a filter creates a delay, but thats just a thing you have to learn to live with. Also, you only get in one sample at a time, which is not enough for filtering, so you have to store the old input values somehow. This is done using a circular buffer. The buffer is circular, because otherwise youd soon run out of memory. Heres a set of pictures to explain the scheme: The buffer must be at least as long as the filter, but it is practical to set the length to an integer power of 2 (In the above example: 2532), because then you can use the binary AND operation to handle pointer wrapping always after increasing or decreasing one (In the above example, AND with 31). Even better, use byte or word instructions, and wrapping will be automatically handled in overunderflows caused by the natural limits of byte or word. Note that the buffer should be filled with zeroes before starting. A similar circular buffer scheme is also often the best solution for implementing the output part of an IIR filter, no matter how the input part was realized. There are both positive and negative frequencies. Until now we havent had to know this, because we have been able to do all the calculations by using sines as frequencies. Dont be fooled that positive frequencies would be sines, and negative ones something else, because that is not the case. In all real (meaning, not complex) signals, positive and negative frequencies are equal, whereas in a complex signal the positive and negative frequencies dont depend on each other. A single sinewave (real) consists of a positive and a negative frequency. So any sine frequency could be expressed as a sum of its positive and negative component. A single, positive or negative, frequency is: and could also be written as: As stated, a sinewave consists of a positive and a negative frequency component. Heres the proof: (The phase of the negative frequency must also be inverted, because it rotates to the other direction) As you see, the imaginary parts nullify each others, and all that remains is the real part, the sine wave. Amplitude of the sine wave is the sum of the amplitudes of the positive and the negative frequency component (which are the same). This also proves that in any real signal, positive and negative frequencies are equal, because a real signal can be constructed of sine waves. The complex Z-plane is a good place to look at positive and negative frequencies: Positive frequencies are on the upper half of the circle and negative frequencies on the lower half. They meet at angles 0 and the Nyquist frequency. Aliasing usually means that when you try to create a sine wave of a frequency greater than the Nyquist frequency, you get another frequency below the Nyquist frequency as result. The new frequency looks like as if the original frequency would have reflected around the Nyquist frequency. Heres an example: The cause of aliasing can be easily explained with positive and negative frequencies. The positive component of the sine wave actually gets over the Nyquist frequency, but as it follows the unit circle, it ends up on the side of negative frequencies And, for the same reasons, the negative component arrives on the side of positive frequencies: The result is a sine wave, of frequency SR-f. Analytic signal It is sometimes needed to first create a version of the original signal that only contains the positive frequencies. A signal like that is called an analytic signal, and it is complex. How does one get rid of the negative frequencies Through filtering It is possible to do the job with an IIR filter that doesnt follow the conjugate-pair-poles-and-zeros rule, but a FIR filter is significantly easier to create. Well use the old formula that we first used to create sinc: but this time, instead of cosines, only including the positive frequencies: As you see, the filter coefficients are complex. We should also halve the amplitude of the positive frequency (it should be half of the amplitude of the cosine, because the negative component is gone) but thats not necessary, because itd only scale the magnitude. To convert the complex analytic signal back to real, just throw away the imaginary parts and all the frequencies get a conjugate (on the z-plane) pair frequency. Here the amplitudes drop to half, but as we skipped the halving in the filtering phase, it is only welcome. The real to analytic signal conversion could also be a good spot for filtering the signal in other ways, because you can combine other filters with the negative frequency removal filter. Amplitude modulation Amplitude modulation means multiplying two signals. All samplepoints in the modulated signal are multiplied by the corresponding samplepoints in the modulator signal. Heres an example: What happens if we modulate a signal with a sinewave The original signal is (as we have learned) a sum of frequecy components, sinewaves of various frequencies, amplitudes and phases. Note that the signal we are talking about here is real, not complex. Say sNUMBER is one of the frequency components. So, we can write the original signal as: Now, if we multiply this signal with the modulator signal m, we get: This is good, because as you see, its the same as if the frequency components were processed separately, so we can also look at what happens to each frequency component separately. A frequency component can be written as: where amp is the amplitude, f the frequency and a the phase. The modulator sine can be written the same way (Only added the letter m): Multiply those and you get: If we discard the phase and amplitude information, we get: which is two frequencies instead of the origial one. Heres a graph that shows how the frequencies get shifted and copied. The original frequency is on the - axis and the resulting frequencyfrequencies on the axis: In the graph Modulated, the frequencies that would seem to go below zero, get aliased and therefore reflect back to above zero. In sampled signal, the Nyquist frequency also mirrors the frequencies. Frequency shifting With some tweaking and limitations, you could make a frequency shifter by using sinewave modulation, but theres a better way. Lets try modulating the signal with e(i mf x) instead of cos(mf x). Phases and amplitudes are irrelevant, so ive just ignored them. (I hope you dont mind) Lets see what happens to a single positivenegative frequency when it is modulated: The answer is very simple. The original frequency got shifted by the modulator frequency. Notice how the rule Multiplication in the time domain is convolution in the frequency domain. applies here also. Heres an example on the z-plane unit circle. p0, p1, p2 are the positive frequencies and n0, n1, n2 their negative conjugate frequencies. Say the modulator frequency rotates the frequencies 14 full cycle counterclockwise: In the modulated signal, the original pair frequencies (like p0 and n0) are no longer conjugate pairs. Thats bad. Another bad thing is that negative frequencies get on the side of positive frequencies and vice versa. But if we first filter all the negative, and those of the positive frequencies that would arrive on the wrong side of the cirle, and then modulate the filtered signal: (The filter formula is in the chapter A collection of FIR filters in section Combined negative frequency removal and bandpass) Now it looks better To make this filtered amp modulated complex signal back to real again, just discard the imaginary part and all the frequencies get a conjugate pair: For most sounds, frequency shifing doesnt do a very good job, because they consist of a fundamental frequency and its harmonics. Harmonic frequencies are integer multiples of the fundamental frequency. After you have shift all these frequencies by the same constant frequency, they no longer are harmonics of the fundamental frequency. There are ways to do scaling instead of shifting, but just scaling the frequencies would be same as resampling, and resampling also stretches the sound in time, so it has to be something smarter. The main idea is to divide the sound into narrow frequency bands and to shiftscale them separately. OK, so frequencies usually come with harmonics - Why Just think where sounds in nature originate from: vocal cords in our throat, quitar strings, air inside a flute. All vibrating objects, and you have probably learned at school that objects have several frequencies in which they like to vibrate, and those frequencies are harmonics of some frequency. What happens in those objects is that they get energy from somewhere (moving air, players fingers, air turbulence), which starts all kinds of vibrationsfrequencies to travel in them. When the frequencies get reflected, or say, go around a church bell, they meet other copies of themselves. If the copies are in the same phase when they meet, they amplify each other. In the opposite phases they attenuate each other. Soon, only few frequencies remain, and these frequencies are all harmonics of same frequency. Like so often in physics, this is just a simplified model. A note about notation. ) The fundamental frequency itself is called the 1st harmonic, fundamental2 the 2nd, fundamental3 the 3rd, and so on. Chromatic scale In music, harmonics play a very important role. The chromatic scale, used in most western music, is divided into octaves, and each octave is divided into 12 notes. The step between two adjanced notes is called a halftone. A halftone is divided into hundred cents. An octave up (12 halftones) means doubling the frequency, an octave down (-12 halftones) means halving it. If we look at all the notes defined in the chromatic scale on a logarithmic frequency scale, we note that they are evenly located. This means that the ratio between the frequencies of any two adjacent notes is a constant. The definition of octave causes that constant12 2, so constant 2(112) 1.059463. If you know the frequency of a note and want the frequency of the note n halftones up (Use negative n to go downwards) from it, the new frequency is 2(n12) times the old frequency. If you want to go n octaves up, multiply by 2n. But why 12 notes per octave As said, harmonics are important, so it would be a good thing to have a scale where you can form harmonics. Lets see how well the chromatic scale can represent harmonics. The first harmonic is at the note itself: 0 halfnotes 1. The second harmonic is at 1 octave 2. The third harmonic is very close to 1 octave 7 halftones 19 halftones 2(1912) 2.996614. E assim por diante. Heres a table that shows how and how well harmonics can be constructed: Not bad at all The lowest harmonics are the most important, and as you see, the errors with them are tiny. I also tried this with other numbers than 12, but 12 was clearly the best of those below 30. So, the ancient Chinese did a very good choice The above table could also be used as reference when tuning an instrument, for example a piano (bad example - no digital tuning in pianos), to play some keys and chords more beautifully, by forcing some notes to be exact harmonics of some other notes. A common agreement is that one of the notes, middle-a, is defined to be at 440Hz. This is just to ensure that different instruments are in tune. Flanger is simply: where d is the length of the variable delay. D values have a lower limit, and the variation comes from sine: Because d is not integer, we must interpolate. Most probably, annoying high frequency hissing still appears. It can be reduced by lowpass filtering the delayed signal. Wavetable synthesis means that the instruments being played are constructed of sampled sound data. MOD music is a well-known example. Also most of the basic home synthesizers use wavetable synthesis. Say you have a sampled instrument, and want to play it at frequency f 440Hz, which is middle A in the chromatic scale. To be able to do this, you need to know A) the samplerate of the sample and the frequency of the sampled instrument, or B) the wavelength of the instrument expressed as number of samples (doesnt have to be integer). So you decide to precalculate the wavelength to speed up the realtime routines a little: The samplerate of your mixing system, SR, is 44100Hz. Now that you know this, you can calculate the new wavelength, the one you want (number of samples): In the mixer innerloop, a sample offset variable is used in pointing to the sampledata. Every time a value is read from the sampledata and output for further mixing, sample offset is advanced by adding variable A to it. Now we must define A so that ol (256) is stretched (here shortened) to nl (100.22727), in other words, so that for ol samplepoints in the sampledata, you produce nl output values: Everything on one line: Thats it By using A as the addvalue, you get the right tone. Click removal There are some situations when unwanted clicks appear in the output sound of a simple wavetable synthesizer: Abrupt volume (or panningbalance) changes. A sample starts to play and it doesnt start from zero amplitude. A sample is played to the end and it doesnt end at zero amplitude. (Biased sampledata or badly cut out sample) A sample is killed abruptly, mostly happens when new notes kill the old ones. Poor loops in a sample. And what does help Heres some advice: Volume changes must be smoothed, maybe ramped, so that itll always take a short time for the new volume to replace the old. Clicky sample starts can be muffled, meaning that the volume is first set to zero and then slided up. This could of course be done beforehand too, and some think muffling sample starts is wrong, because the click may be deliberate. Some drum sounds lose a lot of their power when the starts are muffled. Another case is when the playing of a sample is not started from its beginning. That will most probably cause a click, but muffling is not the only aid - starting to play from the nearest zero crossing also helps. Abrupt sample ends should also be faded down. This may require some sort of prediction, if you want to fade down the sound before its ran over by another sound. This prediction can be made by using a short information delay buffer. It may be easier to just use more channels, to allow the new sound to start while the other one is being faded out in the background, on another channel. When the sampledata ends at a value other than zero, the cause may be that the sampledata is not centered around the zero level, or that the creator of the sample has just cut the end of the sample away. The easiest way to fix this is to fade out the end of the sample beforehand. However, this is not always possible. Symmetric form Turning an IIR filter backwards Getting rid of output(tn) Getting rid of input(tn) FIR frequency response IIR frequency response. Olli wrote he tried to make his text as down-to-earth as possible. Well, heres a more mathematical approach. But Ive still tried to make this intuitive and FUN rather than boring myself with lengthy proofs. This also means that there may be errors, most probably in signs. Symmetric form Say you have this IIR filter: You can put its equation to this symmetric form: Now define a new function, middle(t): You can rewrite this as: Notice how the transition from input(t) to middle(t) is a FIR filter and the transition from output(t) to middle(t) is another. So the IIR filter in fact consists of two FIR filters facing each other. This gives a simple approach to frequency response calculations (see the section IIR frequency response). Turning an IIR filter backwards You can solve input(t) from the IIR equation: Now swap input and output and you have a filter that undoes what the original did. But if the frequency response of the original filter was ZERO for some frequency, the inverted one will amplify that frequency INFINITELY. This is just logical. The inverted filter will also have an opposite phase shift, so that if R(f) is the frequency response of the original filter as a complex number and r(f) is the frequency response of the inverted filter, R(f)r(f)1 for every f. Getting rid of output(tn) Say you have somehow found that you need an IIR filter like this: You need to know both output(t2) and output(t-2) to be able to compute output(t). Doesnt seem very practical. But you can shuffle the equation a little: Now define a new variable ut2 and use it instead of t: Then solve output(u): Now you can use the filter. Getting rid of input(tn) Notice how in the previous example, input(t) became input(u-2). Had there been input(t1), it would have become input(u-1) which can be used in real time filters. Generally, you can get rid of input(tn) this way if the equation also uses output(tm) where mn, because you can define utm which turns input(tn) to input(u-(m-n)) which you get in time. If mltn, this is not possible: Here m0 and n1, so you cant get rid of input(t1) and keep the filter mathematically equivalent to the original. However, you can delay the output by one time unit: Usually, this small delay doesnt matter. But it changes the phase frequency response of the filter and this DOES matter if you then mix the filtered signal with the original one or others derived from it in that case, youd better make sure that all of the signals have the same delay. (Except if you happen to like the extra effect.) (For example, if you have a filter output(t)input(t-1), it doesnt do much as such. But if you mix the filtered signal with the original one, the mixing becomes a filter in itself and you can compute its frequency response and all.) If you try to force the original filter through the utm trick by introducing a dummy 0output(t1) term: youll just get division by zero. FIR frequency response Treat a sine wave as a rotating phasor e(it2piffs) where: The real component of this phasor is the regular sine wave. The neat thing about this is that you can multiply it with various complex numbers to scale the magnitude and shift the phase at the same time. By defining ze(i2piffs), the phasor can be written as zt. This is the same z that is used in pole-zero calculations (see chapter IIR filter design using pole-zero method). Heres the general FIR equation: Now, lets look what the filter does to an infinitely long sine wave with frequency f. But this sine wave can be replaced with the rotating phasor if we then throw away the imaginary component of the output. m(k) is real so the real and imaginary components cant affect each other. Here the zt factor doesnt depend on k, so it can be moved outside the sum: z depends on f (ze(i2piffs), remember) but the value of the sum doesnt depend on t. Ill call it R(f): output(t) is a rotating phasor at the same frequency as input(t) it just has a different amplitude and phase as defined by R(f). This means that for an infinitely long sine wave of frequency f, R(f) shows how the filter affects its amplitude and phase. In other words, R(f) is the frequency response of the filter. Its a complex function. If you dont remember what this means, see section Complex math with filters in chapter Whats a filter in this file. IIR frequency response When two filters are concatenated so that one filters output is fed to the other filters input, the responses are multiplied at each frequency: A filter that just connects its input to its output and doesnt change the signal at all has a frequency response of 1 at all frequencies: Now assume that we have a filter with frequency response R(f) and we make another filter with frequency response Rinv(f) that UNDOES everything the first filter did to the signal when they are concatenated. So the inverse filter also has an inverse frequency response. Remember, an IIR filter consists of two FIR filters facing each other (see section Symmetric form). This setup can be treated as a normal FIR filter followed by an inverted FIR filter: This means that if you can calculate the frequency responses of the two FIR filters, you can calculate the IIR frequency response by dividing one with the other. An example. You have this IIR filter. Change the names of functions a little: Compute the frequency response of filter input1-gtoutput1 (originally input-gtmiddle). The general formulas: In this particular case: The input2-gtoutput2 (originally output-gtmiddle) filter: Now the whole IIR: To actually calculate the frequency response at some frequency, youd apply Eulers formula and the usual complex number rules: R in the filters means resonance, steepness and narrowness. Fastest and simplest lowpass ever Fast lowpass with resonance v1 19 Comments raquo Thanks for posting this. It8217s a nice collection of audio DSP nuggets. May I suggest that the URL at the top of the original text document (iki. fiodspdspstuff. txt ) be pointed directly to this page. Comment by ColdCold 8212 2009-11-16 16:06 Thanks Mate, Greatly appreciate this tutorial. DSP in simple terms is not easy to come by on the Web Comment by Don 8212 2010-05-10 04:29 Thanks a lot. Very useful concepts explained in a lucid manner. Comment by Ravi 8212 2010-08-30 14:59 Hi, About notch filter.. Why I can8217t get the frequency cut effect Sample rate:1600 freq 1950 q 0.1 z1x cos(2pifreqsamplerate) a0a2 ((1-q)(1-q))(2(fabs(z1x)1)) q a1 -2z1xa0a2 b1 2z1xq b2 -(qq) 8212821282128212- frequency: 1950.000000 q: 0.100000 z1x: 0.195090 a0a2: 0.438887 a1: -0.171245 b1: 0.039018 b2: -0.010000 Each sample calculation: 82128212821282128212821282128212821282128212821282128212 reg0 a0a2 ((double)samplecurrentsampleminus2) a1sampleminus1 b1reg1 b2reg2 reg2 reg1 reg1 reg0 82128212821282128212821282128212821282128212821282128212 Is it correct Output is clean voice, but 1950Hz carrier is still there. BR Comment by Alexander Vangelov 8212 2011-03-16 22:46 Freq should be between 0 and samplerate2. (Just a quick comment before I go to bed) Comment by Olli Niemitalo 8212 2011-03-17 00:53 Thank you, it works :) I missed a zerro digit in parametters (just before I go to bed) Sample rate: 16000 Freq: 1950.000000 q: 0.400000 z1x: 0.720854 a0a2: 0.504599 a1: -0.727484 b1: -0.576683 b2: 0.160000 Comment by Alexander Vangelov 8212 2011-03-17 10:43 Very good tutorial, thanks Comment by Vadim 8212 2011-10-11 19:42 man, this is the best introduction (covering all topics) into DSP I stumbled upon perhaps I do have a chance to pass the exam. D sorry, for a double post. but8230 can you attest everything is correct for example, 822082218221 You can use any number of poles and zeros, but they must all have 8220conjugate pairs8221, if they are not positioned on the 8220-8221 axis. 822082218221 is this true I8217m playing with applets that allow for poles without conjugate pairs and seemingly band-pass filters (with regard to the magnitude response) can be built this way. can you please explain ( laps. fri. uni-lj. sidpsarhivappletiisipsystemv4.0srcapplet. html ) Doug, it is true, IF you want the filter to have a real output, not complex. If you make a bandpass with just one pole, and have the pole so close to the unit circle that the filter output is pretty much a single frequency, then the output of the filter will be a complex phasor rotating in one direction on the complex plane. If you switch the sign of the imaginary part of the position of the pole then you get as output a phasor that rotates in the opposite direction. If you have poles in both of those positions, then the output must contain both of those complex phasors in equal parts, thus the imaginary parts of the phasors cancel each other. So you get as output a real sinusoid. Good luck with the exam Comment by Olli Niemitalo 8212 2011-12-27 13:41 This is the first cogent explanation of poles and zeros that I have ever received. I feel better and worse at the same time, if you know what I mean. Em qualquer caso. THANK YOU Comment by Mark McConnell 8212 2012-05-09 01:12 8230 Yehar8217s Digital Sound Processing Tutorial for the Braindead 8230 Nice Job Men82308230. I found it very helpful. obrigado. Can you put implementation of audio effects in computer. Comment by Trnform3r 8212 2012-09-16 10:07 Sure, for example as a VST effect. Comment by Olli Niemitalo 8212 2012-09-16 22:14 This is fantastic nice work and a very well explanation of DSP. Thank you :D Comment by tor 8212 2013-02-16 01:42 Thank you so much for this informative writing on the subject which makes life much easier since no-where could I find any book on the subject which makes it as clear as you did here. Keep it going and thank you again. Comment by FJ Botha 8212 2015-02-21 10:14 Frickin delicious Seriously, i thank people like you for simply existing and count my blessings that i found this brilliant introduction you created. The note takingoutline is digestable in one bite and it will stick with me during my upcoming solo winter sound holiday to the pampa and magellians strait, the large uninhabited Falkland rock, and if im still alive - christmas island. Dec to Feb. I hope to capture enough sound to keep me glazed and deadeyed until black metal villians capture Oslo Comment by Mick Dkaye 8212 2016-10-18 19:13 And love that Black Deck. Masonna weeps Comment by Mick Dkaye 8212 2016-10-18 19:16 Leave a commentDSP filter - DISPFIL. EXE Version 1.09J - June 18, 2000 JE3HHT Makoto Mori Translated into English by JA7UDE Oba This is a DSP filter tool using a PC with the soundcard. With this tool, you even can design various types of digital filters including adaptive filters. However, this tool is just experimental and will not afford the practical use for amateur ham radio. You probably need a powerful CPU to make this tool run flawlessly. In addition, you need a soundcard with the full-duplex mode. I made this program just because of my own interest. As I was not quite familiar with the use of the soundcard, it still has substantial time lag from input to output and might not well work for CW. Needless to say, this program is freeware. How to uninstall DSPFIL does nothing to the Windows registry, so just delete all the files with the directory that has DSPFIL files. OS: Windows 95, 98, NT (Note - by VE5KC - has worked fine in tests with Windows XP amp Vista) PC: The faster the better Display: 640 x 480 or more Soundcard: 16-bit soundcard that is capable of FULL-DUPLEX (some cards wont work) Hookup and Operation Connect the speaker out of the radio to the Line-in or Mic of the soundcard. Connect a headphone or speaker to the output of the soundcard. Since the Mic input has too high gain, I recommend the Line-in. Adjust the input level by using Mic or Line level in the Record property or the audio in the control panel. You can do that by using the AF gain of your radio, too. Adjust the output level by using Wave or Master level in the Play property or the audio in the control panel. You also can do that by using Up (up arrow) or Down (down arrow) button on the DSPFIL window. If you have a sound output from your speaker without running DSPFIL. EXE, your PC is configured so as to play the recording signal directly and thus you must turn it off. Go Play property and get Mic-in or Line-in muted. If you hear a sound immediately after starting DSPFIL. EXE, you are ready to go. In case you see a message like quotCannot open the sound device, quot your soundcard probably does not support the full-duplex mode. Give up listening to the filtered sound, but you can observe how DSPFIL. EXE works by the FFT and adaptive filter response windows. Since there is a time lag between the input and output, you should keep the buffer size as small as possible. The time lag has a big trouble in filtering CW signals (you will soon understand what it is when you transmit a signal, Hi). Too high input level causes distortion in the analog circuit of the soundcard. You have to adjust the input level by monitoring the FFT display set to quotIN. quot When overdriven, DSPFIL shows quotOverquot in the upper right corner of the FFT window. When the HPF button is depressed, the 100Hz high-pass filter is activated to the input circuit. It is effective if you have DC ingredient, but it raises the CPU load. Use it only when you need it. Details of the filters This is a comb filter using moving average. This filter, by its structure, gets the actual center frequency Rfo shifted from the defined center frequency Fo by RFo fss int(fssFo) Hz ifss fs 2j This can be compensated by carefully choosing the sampling frequency (fs). However, the sound blaster card does not allow fine tuning around 11025Hz, so DSPFIL admits the shift, Hi. The filter does not use a simple averaging calculation but uses subtractions for 12 periods. Thus, the even harmonics are suppressed, but the odd harmonics can be passed through. It is a good idea to use a 500Hz filter of your radio. It has lower quality in the frequency domain compared with BP100, but I think this filter gives the best performance particularly for weak signals. This is a band-pass filter using an FIR filter. It uses x3 oversampling. The physical sampling frequency is 11025Hz while the application sampling frequency is 3675Hz. If the number of taps is increased, the filter become sharper. However, it increases the processing time at the same time, and therefore it will not run on a slow PC. This is an adaptive band-pass filter for CW. I have not tested a lot on the values of fnofEcirc(mu) and fnofAacute (gamma), but I think the filter works, hi. This filter does not affect Fo or Tap, which is configured in the main window. The frequency-domain graph in the lower right corner shows the frequency characteristics of the transversal filter calculated with the coefficients, which are changed by LMS. You can see how the adaptation is performed by changing the frequency of the input signal. In case of weak signals, the filter coefficients tend to be small, which would result in a low level output. To compensate this, LMSB2 leaves the AGC turned on to increase the volume for the weak signals. This is a fixed frequency BPF for SSB. The low-cut frequency is fixed to 200Hz. If it oversamples the 2.2KHz or higher signals, it causes folding errors because of the decimeter. This filter does not affect the Fo, which is configured in the main window. This is a noise smoother for SSB. The adaptive operation might not be well tuned yet. The SSB signal is smaller autocorrelation than the CW signals, so I put small values in the correlation delays. This filter does not affect the Fo or Tap, which is configured in the main window. This is an automatic notch filter for SSB. It would give better results if it had faster convergence behaviour. However, I dare to focus on the response speed for CW signals. This filter does not affect the Fo or Tap, which is configured in the main window. This is a user-customizable filter. The default setting gives a wide band-pass filter for SSB. You can customize it by pushing the DESIGN button (the button face text is written in Japanese). You can copy the parameters of the selected other filter to those of this filter by pushing COPY (the button face text is written in Japanese) button. This filter does not affect the Fo or Tap, which is configured in the main window. User setting for the adaptive filters LMSBP, LMSNS, LMSAN are built-in filters, but the user can design a LMS filter by himself. Push DESIGN (this text is written in Japanese, so it might not correctly appear with non-Japanese Windows) button and select LMS, then push UPDATE (in Japanese text) button. Now you can change the parameters. The algorithm used in the adaptive filters is called Leaky LMS (Least Mean Square method). The user customizable parameters are: Tap the number of orders of the transversal filter Delay the number of delay nodes 2u the response speed V (gamma) the dumping factor Larger u gives faster response but slower convergence. Smaller V (gamma) makes the coefficients decrease faster when the input signal is cut off. However, too small V (gamma) will result in oscillation. Generally put a value a little bit smaller than 1 to V (gamma). If the REVERSE OUTPUT (in Japanese) is checked, DSPFIL outputs an error signal. It is checked to design an automatic notch filter. When AGC is checked, DSPFIL automatically increase the output volume for weak input signals. The characteristics of the adaptive filters are dependent not only on u (mu) and V (gamma) but also on Delay and Tap. Change all of them to see what happens. Use parameters that are given by another design software If you want to test the filter coefficients that are calculated with another filter design software, try the following steps. 1. Push DESIGN button in User1. User6 2. Select User and push Update 3. Push SAVE button and make a filter definition file. 4. Exit DSPFIL. 5. Edit the filter definition file using a text editor. Sampling frequency No over sampling 11025Hz OverSmp1 Over sampling x2 5512.5Hz OverSmp2 Over sampling x3 3675Hz OverSmp3 Filter order Put the number of orders to TAP field Coefficients Put coefficients to H0. Hn (n Tap) fields 6. Start DSPFIL and push one of User1. User6 7. Push LOAD button to load the definition file. CPU Power Since this program intensively uses floating-point operations and is not well optimized, it will not run on a PC with a slow CPU. 73 de JE3HHT Makoto MoriLoad error in equalizer data in some of the 8.21 builds, due to the 62986299 equalizer frequency fixpressors: Added Feedback2 mode. Different behavior from normal Feedback mode, Ratio is . New product: Thimeo WatchCat. Avant version added. RDS encoder: Added code 0 for FM 87.5 in the AF list. Seems to be supported by some receivers. Stereo Tool: Added info to. sts files to allow importing them in Omnia SST. SST: Added possibility to import Stereo Tool settings. SST: Attack and release are now faster when slider moves to the right. SST: Attack Multiplier: Attack is now faster when slider moves to the right. SST: Improved naming of many things, to be compatible with other Omnia products. SST: Disabled MULTICORE for latency 128 and 256, and LQLL. SST: GUI: Added glowing yellow line around SST tray icon, to make it more visible. Bug fix: ACR Stereo caused some mild crackling. Bug fix: RDS quadrature mode could cause crackles and overshoots. Bug fix: SST: Loading a preset stopped server communication. Some changes could get lost due to this. Bug fix: MAC VST version didnt work in OcenAudio (actually a bug in OcenAudio). Bug fix: SST: Multiband text below horizontal meters was upside down. Bug fix: Restarting on unhandled exceptions didnt work in 64 bit version. Bug fix: VLC: potential crash when switching between streams due to memory allocation issue solved. Bug fix: 1 channel audio processing was broken in 8.20. Bug fix: More-than-2 channel audio processing was broken in 8.20 due to what appears to be a compiler bug. Bug fix (Stereo Tool, SST, Omnia.7. 9. 9sg, possibly .11, BreakawayOne): Time zone changes ignored after startup on Windows (compiler runtime bug). Bug fix: BS412: Singleband compressor was placed after BS412, should be before. Bug fix: SST: Reset settings when loading preset, it kept the current settings if they were not present in the file. Bug fix: UECP client could hang after connection close. Now a new incoming connection kills the existing one. Bug fix: Equalizer frequency could be either 6298 or 6299 due to rounding errors, causing preset load errors. MicroMPX: New license mechanism for ARM version implemented. Code improvement: Less duplicate code in binary for Compress. Smaller binary, faster build, better performance. Added option to run Stereo Tool as a service. RDS: Moved reading RDS from file to a separate thread, should avoid hiccups when reading from network drives. Added code to handle (restart) and log crashes in sound card drivers and VLC. Not yet fully tested. Linux: Now using Ubuntu 14.04 instead of 16.04 to build (better backward compatibility). Linux: Now statically linking libstd (better backward compatibility). Linux: Switched to newer compiler for better performance (20, on both Intel and ARM). Web server: Whitelist allows as in 192.168.. instead of 16 (Leifs formatting), always allow 127.0.0.1. Compressors: Volume jumps on preset changes reduced (compressor adjusts to changing target level). Declipper: Detection (veil) didnt always work properly with gt 60 kHz input sample rate. Performance: Reduced CPU load of IO and processing framework. GUI: Hovering over tray icon now shows version number. GUI: Changed scale of many sliders, dB scale sliders are now flat over the whole range. Bug fix: Potential sound card glitches (bug introduced in 8.12). Bug fix: Crash reported by Camclone Bug fix: FM test tones were broken when using Ignore high frequencies (bug introduced in 8.12). Bug fix: VST plugin no longer cuts off the start of songs (bug introduced in 8.14). Bug fix: UECP port was always used (even if FM disabled). Bug fix: Potential bug in non-ASIO code could cause crashes depending on compiler optimizations (never seen). Bug fix: Potential crackle in lowest multiband band fixed. Rewrote error logging code. Omnia 9sg: Builds with the new framework now. SST: Make IPPORT changable without web gui. SST: Preset list was empty on first start SST: Improved naming of settings in Processing section. SST: Web interface code compiles separate from SST now. SST: Added importing Stereo Tool presets. SST: Multiple MicroMPX output IP addresses and ports CN. SST: Launch browser automatically on first start. SST: Improved startup screen. SST: Currently active presets highlighted. SST: Prepared for longer list of MicroMPX IPports (partially there). SST: Clicking on meters in top bar takes you to the page directly. SST: Top bar meter that corresponds with active screen is highlighted. SST: Add glowing yellow line around SST tray icon, to make it more visible. MicroMPX decoder: Buffer underruns seen on ARM version, caused by loggingtracing library. MicroMPX decoder: Shows more startup info and waits for pressing ENTER at end (command line). MicroMPX decoder: Improved handling of out-of-order and lost packets. Added built-in self test for sound cards. SST: Slider behavior improved (slow vs fast moving the mouse). SST: Preset list was sometimes empty, message added that in this view no presets are available yet. SST: Preset list jumped to new view after opening which looked weird. Fixed. SST: Composite Clipper is on by default. SST: Added presets. Bug fix: SST: Rules are made to be broken disabled when not using dual side band. Bug fix: SST: Clip LR peaks above was always disabled. Bug fix: Glitches due to missing thread priorities fixed (since 8.12). Bug fix: Non-2 channel audio (1, 5.1, 7.1 etc) didnt work (since 8.12). Bug fix: Stereo Tool web interface was adding 20 marks in text boxes when typing spaces (since 8.12). Bug fix: ASIO in 192 kHz, non-ASIO Normal output set to input audio got 48 kHz audio causing underruns (8.12). Bug fix: CPU core affinities Main 1 and 2 were overwritten by LQLL affinity settings (since 8.12). Bug fix: Volume drop when using old Calibration settings (since 8.12). Bug fix: Heap corruption on close for command line version (only visible when a debugger was attached). Bug fix: Linux: Web interface exceptions when closing the web interface. Bug fix: Weird crash in Adobe Audition 3.0 (from 2007), gone after changing compiler settings. Possibly bug in Audition or compiler. Bug fix: Separate thread for SCA input with ASIO didnt work, disabled that option now. Bug fix: GUI: ASIO inputs sometimes showed dotted overlay moving from half-way to top (and disappearing). Bug fix: GUI: Values for Max Attack and Max Release were displayed incorrectly. Bug fix: ASIO buffer was too small, could cause glitches after a buffer underrun. Bug fix: Improved sound card synchronization constant clock speed determination. Bug fix: Sound card timeout times increased to take VLC connection time into account. Bug fix: VST version didnt work in Wavosaur. Cause unknown but it works now. GUI: Gate vs Gate Freeze ranges were different, causing sliders to go outside of their range. VST plugin: Process function crashed when called with lt 0 samples (Wavosaur empty file bug). Added, removed and updates presets. Framework: Added support for upto 7(7) threads, works much better on slower CPUs with many cores. Framework: Added Quality settings 100 (OmniaSST only). With this, latency can go down to 128 samples (5 ms analog input to analog output). Framework: Smooth preset switching - no more hiccups if the same filters are used, also faster. Linux version: High CPU load (100) in one thread fixed. Linux version: Made plugin for Rivendell (and other programs that want to use it). Not yet fully tested. Linux version: Didnt run on Fedora. Fixed. Memory usage reduced. Sound cards: Resampler improved, now measures separate buffer filling and speed offsets. AGC: Added Gain slider. AGC: Added Windowing. Compressors: Added Windowing. Composite Clipper: Rules are there to be broken mode could cause occasional 1-sample overshoots. Fixed. Low Latency monitoring: Affinity setting added. Low Latency monitoring: Latency selection added. Removed lower lowpass filter for high frequency input at lower latencies, not needed anymore. GUI: Showing AGC gating the same way as for compressors GUI: Low Latency monitoring: CPU usage display added. GUI: Added Hear buttons to listen to the output upto this point. GUI: Fixed synchronization issue between meters, scopes and audio. GUI: Help pop-ups also work on Mac OS X now. GUI: Renamed PNR Noise Hum to Dehummer. Web server: Multi-threading added. Now handles upto 10 requests simultaneously. Web server: no-cache added to objects. js. Web server: Automatically add index. html if a directory is specified. Web interface: Added white list for logging in. Web interface: Added spectrum display for Delossifier and Absolute Highs. Web interface: Buttons didnt work. Generic plugin: Added option to make the plugin access input and output sound cards (Windows only). Bug fix: Potential crash in ASIO when changing channels fixed. Bug fix: Potential crash when zooming waveforms fixed. Bug fix: Latency with ASIO was not fully predictable on restarts. Fixed, allows for slightly lower latencies. Added Delossifier, repairs MPEG-compressed audio. Added Absolute Highs, generates lost (lowpass filtered) high frequencies. Added FM MPX Analyzer with MPX demodulator, RF bandwidth view and LeftRightQuadrature () decoder. Added RDS ASCII external input support for programs such as Arctic Palm, RDS Magic. AGC: Added Ratio control. Composite Clipper: Added Rules are there to be broken mode for extra loudness and clarity. Clipper: Added Multipath Stereo Phase Difference Remover to improves reception and reduce multipath. FM: Added Polar stereo encoder for OIRT 65-74 MHz (Soviet system) FM stations. FM: BS412: Filter bass based on ITU.1770 instead of the existing kinda-randomly-chosen weighing. AM: Added test tones, including a sawtooth with a pauze at 0. Improved resampling between sound cards (more improvements are still needed). Raspberry Pi: Optimized the code a bit. Linux versions: Remove wxWidgets, replaced by pure XLib. Mac OS X versions: Remove wxWidgets, replaced by pure Cocoa (VST: only 64 bit so far). GUI: Auto-hiding deprecated pages and sections if they are disabled: AGC: Compatibility amp behavior header Bass AGC (Old) Classic Multiband Compressor Classic Singlenband Pre-limiter (no longer needed since Leifs clipper improvements) Advanced Clipper: Bass protection (Deprecated) heading Advanced Clipper: Highs gap protection heading (replaced by Leifs gap protection) GUI: Auto-hiding deprecated settings if they are set to the optimal value: Leifs clipper efficiency modification checkbox (overruled by Leifs clipping strictness) Composite clipping Strictness (CPU) checkbox Phase rotation: At start of processing checkbox All Singleband and Multiband Compressor type (Analogdigital) pull-downs Multiband and Natural Dynamics Flat Frequency Response Compatibility mode (bad) checkbox Advanced Clipper: Smooth slope setting (under Distortion) Advanced Clipper: Upsampled highs clipping (useless) setting (under Distortion) Advanced Clipper: Stokkemask: Force Stokkemask even if not using Composi te Clipper (bad for audio) BS412: De-basser non-ITU.1770 compatibility mode (bad) checkbox GUI: Added tool tips. GUI: Add indicator that a setting has been modified. GUI: Added reset mode to last-loaded setting beside the existing reset to default. GUI: Made scope skinnable. GUI: BS412 disabled Advanced Clipper Drive. GUI: AM: Added Forcibly remove DC offset to the AM settings panel. GUI: Hide Stereo Tool and Thimeo names in whitelabel builds. GUI: Made distinction between switches and onoff buttons. GUI: Added Multiband Quick Overview screen with the most important settings. GUI: Redesigned Simple mode pages: Single screen overview. GUI: Added names Diversity Delay and HD to indicate that it can be used for HD. GUI: Made R3LAY skinned version for LAWO R3LAY. Command line version now also handles licensing properly. Bug fix: Fixed VST DLL unload bug that affected Adobe Audition. Bug fix: Fixed crackle at startup caused by Ignore High Frequencies (even if disabled). Bug fix: BS412 was still active when FM was disabled. Bug fix: Sound card setting Input without processing didnt work in stand alone version. Bug fix: ASIO: Fixed 1624 bit LSB modes used by new AudioScience 192 kHz card. Bug fix: ASIO: Added a check to get rid of a small memory leak in the ASIO SDK. Web interface: Added Access-Control-Allow-Origin: to binary HTTP protocol to support cross-server scripting. Generic plugin API: Added GetApiVersion function. Generic plugin API: Added stereoToolIsLicenseValid function. Added, updated, removed and reordered presets. Improved quality of bass, ringing artifacts cleanup filter caused some clicking effects in low bass (Since v 5.00). Added 8x oversampled non-phase linear Butterworth de-emphasis to match some hardware units pre-emphasis. Sound cards: Added backup sound card support (sound card 1 switches to 2 after some silence). True Bass: Add highpass filter for lower latencies because it can cause very low subs, caussing problems on FM. Highpass filter: Improved non-phase linear HPF a bit, works better when used after other filters now. ASIO: Only claim the channels that are actually in use. ASIO: Add already in use as possible cause to ASIO claimBuffers error message. ASIO: Added LeftRight indications in dropdown list. ASIO: Changed default channels to disabled except for Normal Output and Input. ASIO: Add more output selections - Increased from 8 to 32. Ignore High Freqs: Added Auto frequency option. Determines lowest possible value for max CPU load reduction. PNR: Settings now stored in. sts file for VST plugins (partial fix of the problem). Clipper: De-esser: Changed default value, now disabled by default. Reduces the CPU load. GUI: MPX display disabled when waveform display is disabled. GUI: Improved RESET. now gives preferred values instead of traditonal values needed for old preset compatibility. GUI: Disable Composite Limiter slider when Hard Limit is disabled. GUI: Bug fix: Phase Delay was not grayed out in Bypass mode. Bug fix: Some settings were net reset when loading a default preset in previous version. Bug fix: VST: GUI-related hiccups in Adobe Audition. Bug fix: Synchronization to external RDS signal could sometimes have a hiccup due to a threading issue. Bug fix: BS412 level calculation was wrong for LR instead of MPX output when using non-Pre-emph output. Generic Plugin: Add function to Get parameter values. Added new presets. FM: Add extra pre-emphasis for clipping for stations that send the processed FM signal through a lossy codec. Web interface: Added Presets dropdown menu and Bypass in web interface top bar. Sound card control: Slightly better low latency handling for non-ASIO. Sound card control: Auto synchronization mode means OFF when any of the affected buffer sizes is lt 250 ms. Bug fix: Error in AM AF frequency list for RDS. Some high AM frequencies might not have been encoded correctly. Bug fix: Simple Clipper Pre-limiter amplification was performed even when Simple Clipper was disabled. GUI: Bug fix: True Bass was called WARM BASS in the LOADSAVE menu, and WB in bottom menu. GUI: Bug fix: Lowpass filter frequency was displayed incorrectly for Separately processed streaming output. GUI: Bug fix: Pre-emphasis for MP3 was shown disabled when using Separately processed streaming output. Code: Redesigned for faster compiling. Updated presets. Generic plugin: Added function to set parameter values. Generic plugin: Added export of ParameterEnum for generic plugin customers. Bug fix: In 7.80 Automatic restart could be triggered incorrectly when the buffer size was very small. Bug fix: Fixed some small issues caused by the Thimeo framework rewrite, had no effect. FM: Improved Stokkemask (ITU-R. SM1268) filter for composite clipper (sounds better). FM: Fixed several bugs, mainly noticeable in SSB mode, at lower Quality settings. FM: SCA1 channel is now displayed in MPX graph. Updated and added some presets. FM: Added composite clipper FM: Added SSB (Single Side Band) mode. (Requires composite clipper.) FM: Added Multipath clipper. (Experimental Requires composite clipper.) Loudness: Improved Airy Highs. Highs now sound crisper and less tonal. Loudness: Rewrote clipper to reduce CPU load (Strictness can be set lower to give similar results). Added Phase equalizer. Updated and added many presets. User interface: 5 modes ranging from Simple to Extreme Tweaker. Basic should be good for most people. Greatly reduced CPU load of almost all filters. With identical settings, the CPU load is typically 12 reduced. Singleband compressor completely redesigned. Keywords: Attack, release, release hold, ratio, feed forward vs feedback, lookahead, burst protection. Multiband compressor completely redesigned: Configurable number of bands monitoring individual bands, median display. Added graphics equalizer. AGC: Added side chain. AGC: Added singleband compressor after AGC. AGC: Smoothed Band 1 to band 2 coupling. AGC should work better if declipper or noise gate is activated. Extra lowpass filter before processing to fix problems with songs with excessive very high frequency content. Set 0.25 kHz above actual LPF freq, but never below 16 kHz. Loudness: Improved delayed bass clipping protection to cause less distortion. Provision to discard buffer after a certain period of inactivity added for VST plugin. FM: SCA was missing since version 7.00, added again. Declipper: Reduced artifacts when using declipper in combination with non-phase linear highpass filter. Declipper: Changed default settings based on input from Jesse Graffam. Sound cards: Synchronize to output now also works with Normal output. User interface: Made 6 dB lines in meters more prominent. Bug fix: Disabled mouse hold behavior for all widgets except sliders. Updated and added many presets. Loudness: Added delayed bass clipping for better punch. Loudness: Fixed Bass Shape artifacts, and now also works on higher frequencies. Loudness: Added option to dynamically switch off distorted midshighs for tones. RDS: Better special character display in GUI. Updated presets. Added 5.1 and 7.1 audio support (channels are declipped, highpass and lowpass filtered and clipped, nothing else yet). Loudness: Improved bass clipper, better punch and warmer sound with more mid-bass. (old vs new) Loudness: Removed volume drop between 2.8 and 3 kHz. RDS: Added convertor from Windows-1252 to RDS characters to support special characters. Installer made smaller by moving non-SSE2 version to a separate installer. Now below 1 MB again :) Bug fix: Drawing issue in 7.00 and 7.01 pull-down menus without scroll bar when using mouse wheel. Bug fix: Left and right channels were sometimes swapped when loading a preset in 7.01. Bug fix: Stokkemask clipper was broken in 7.01. Carbon Coder plugin now supports R128 normalization for 5.1 and 7.1 audio. Added some presets. Solved several VST GUI issues in several VST hosts. Made VST window a bit bigger. On loading built-in presets, most FM and sound card settings are no longer reset. Volume increase on preset reload should not occur anymore, although Im still getting reports that it does. But at least one cause is solved. Added sound card restart button, was missing in version 7.00. Bug fix: GUI unresponsive at very high CPU loads when you click right of text in text box. Bug fix: BS412 acted weird with FM sound card enabled, pre-emphasize output disabled at low input sample rates. Updated presets. Total GUI redesign. Interface dynamically adapts to screen size and is nearly touch screen ready (keyboard needed). Added new version alerts and specific warnings which unregistered functions are in use. Last selected preset name visible after restart. Added web interface (no access protections yet). Completely rewritten parameter handling code solved bugs and reduced the chance of introducing new ones. Added ability to reset, load and save parts of presets. Added input sound card Gain control. FM transmitter audio synchronization added to synchronize streaming audio at multiple transmnitter sites (experimental). Loudness: Changed bass clipper behavior, flatter frequency response sounds a bit warmer. Loudness: Bass clipping filter bug fixes: Whistle artifacts gone, knock-on-wood artifacts reduced. Loudness: Airy Highs turned off if there are very little highs (reduces crackling sound that sometimes occurs). Declipper: Reduced clicking artifact in reconstructed audio. Declipper: Improved splitting load over 2 cores. FM test tones: Added Bessel null frequencies and smooth square wave (sounds less loud) test tone mode. Removed a number of memory allocationsfrees that occurred during processing. Added a lot of presets. Natural Dynamics is available in the GUI again, if you enable it explicitly in the INI file. Added sound card input and output tilt adjustment (2nd order). Loudness: Changed clipper distortion masking model, giving less distortion at nearly the same highs level. Loudness: Added Airy highs, makes highs sound both louder and more natural (less restrained). Loudness: Removed bugs that caused hole punching by loud highs. Loudness: Highs Priority no longer causes volume drops caused by loud highs. Loudness: Redesigned de-esser for pre-emphasized sound. Bass sensitivity: If there are both mids and highs, the drop is a bit less strong than before. Bass sensitivity: Determining more precisely when the clip is needed. BS412: Added max deviation (- 75 kHz) display. BS412: Added headroom slider. Determines margin for error to avoid sudden volume drops. BS412: Max deviation can be lowered to create denser sound and protect against sound card problems. BS412: Faster rise after fast drop for louder end result. GUI: Added displays so you can see what is being sent out to the sound card (including calibrations). GUI: Cleaned up Loudness panel (removed some things that are no longer needed). Bug fix: Latency 512 didnt work properly. Bug fix: Fixed GUI issue where singleband compressor settings were not always correctly loaded. Bug fix: AXIA input sound card loading issue. Added Carbon Coder plugin with multi-pass R128 support. AGC: Added a slider to protect against volume drops for loud voices. Loudness: Improved overall filtering (removed post ringing) at latency 4096, sound is cleaner in many ways (especially bass). Added some new controls for configuring it (Time Spread) Loudness: Advanced Bass Distortion Protection: Improved bass clipper. Bass is now louder and clipped tighter Loudness: Added bass shaping. Adds harmonics to very low frequencies to make them sound louder Loudness: Allow turning highs intermodulation distortion off Loudness: Added highs priority slider, increasing this reduces high frequency intermodulation distortion but may cause some drops in other frequencies Loudness: Asymmetric bass reduction should not go below dynamic threshold Loudness: Reduced strictness, the high precision wasnt really useful Loudness: Split sensitivity for bass clipper: Can now be configured separately for mids and highs. Result: more constant and higher bass level Loudness: Turned off old bass protection mechanisms in default settings. Caused artifacts, no longer needed Loudness: Reduced clipping strictness for mid-bass frequencies (lower CPU load) Loudness: Increased de-esser range, and it can be turned off now Loudness: Simplified clipping algorithm Stereo Boost disabled for lowest frequencies (below approx. 140 Hz) BS412 limiter: Show a warning and info when BS412 is enabled BS412 limiter: Fixed drops issue. Very brief drops can still occur but they are usually hardly noticeable BS412 limiter: Changed default values for more constant sound level Added and updated many presets Split values for Singleband, Singleband AGC style and BS412 Bug fix: Stereo Image was broken for latency 512 Bug fix: Icon now visible in stand alone version Bug fix: Stokkemask clipper didnt work at latency 512.Major code cleanup and restructuring Added BS412 limiter Added Stokkemask clipper Improved scope display (triggered) Improved low latency (6 ms) monitoring output in Stand Alone version Improved overall quality by changing the processing window shape for ALL filters Added support for cheaper limited WinampDSP declipper license Reduced CPU load when playing pure silence AGC: Improved handling of sudden loud bursts Loudness: Improved protection of mids against loud highs, which were causing drops (holes) in the sound. Loudness: Changed the bass protection shape Loudness: Added protection against distortion caused by asymmetric bass sounds Loudness: Slightly better protection of frequencies below 700 Hz against very loud bass Loudness: Bass protection and de-esser levels were calculated incorrectly if Final Clipper was enabled with a volume other than 1.00 BS412 limiter now also works with Pre-emphasize output disabled. Phase rotator: Improved behavior and display Bass AGC: Now works on a much smaller frequency range so it only affects really deep bass sounds Bug fix: Changed default values for the Allow more distortion sliders in the VST version Bug fix for Omnia 9: RDS AB flag works properly now, and if you change the RadioText and there was no sequence of texts there the new text will start to be broadcast immediately Bug fix: Forcibly remove DC offset in Loudness did not always work properly Slightly reduced Loudness CPU load, but the new highs protection adds a lot more if its enabledAdded multicore support for Declipper and Loudness. Throughput is now 38 higher in presets that use both. Loudness: Fixed a bug in the non-SSE2 version. Declipper: Fixed memory bug in clipping detection. This may have lead to weird issues. Improved performance when closingminimizing Stereo Tool if the virus scanner NOD32 is used. Bug fix: Loudness de-esser setting was not saved in VST plugin. Bug fix: Loudness values above 3.00 were not loaded correctly in VST plugin. Bug fix: TA button in main window did not work. Changed a value in Bojchas Neodymium preset. There is now also a separate declipper program See perfectdeclipper. Added declipper filter Loudness: Completely redesigned for more consistency, better transients, louder output and greatly improved highs. Punch is no longer needed. Loudness: Improved vibrating voices filter caused by loud bass. Loudness: New filter that removes highs distortion caused by loud bass, now allows bass levels upto 100 without (much) distortion in the highs. Loudness: New filter that detects when highs cause volume drops in the rest of the sound, and briefly lowers the highs level. Loudness: Added oversampling checkbox for streaming and mastering: This makes sure that no clipping will occur if the audio is upsampled afterwards. Loudness: Added separate dirty bass mids highs sliders for better bass control and artistic effects. AGC: Redesigned to handle different input levels much better. AGC: Enabled ITU-1770 support (for bass and head separately). AGC: Added stereo separation control. AGC: Added start level for mastering. AGC: Separate Remove remaining peaks settings for lows and mid-highs. AGC: Added Force level below slider for improved peak control. Forces the AGC level down faster if the difference gets too big. Added Bass AGC filter to reduce excessive bass sounds. Multiband: Added voice protection settings to better cope with loud vocals in music. Multiband: Allow changing band frequencies (only via STS file). Multiband: Added control of link between bands 1 and 2. Pre-emphasis: Moved to end of processing, just before clipping. Highpass filter: Improved filtering to reach a cleaner output. Highpass filter: Added order setting for non-phase linear filter. Improved audio processing window shape to reduce artifact levels. Improved pre-ringing filtering to avoid artifacts that occurred before. Added SCA output channels (SCA1 works, SCA2 is problematic due to the required frequency range). Support added for multiple Stereo Tool installations with different settings per installed version (requested by Jazler). Added registration to command line version. Bug fix: Fixed Test right channel in FM Transmitter Calibration (broken in version 6.10). Bug fix: FM Calibration frequencies 23-57 kHz were not at the correct frequencies. Bug fix: Fixed leftright channel swap that could occur in the stand alone version. Bug fix: AGC could hang in some cases when enabling more bands. Fixed. Bug fix: Phase rotation did not sound good at low latencies in version 6.10. Bug fix: Time zone in RDS was wrong. GUI improvement: Made waveform background black. RDS: Adjusted behavior if no special characters (lt, gt, , ) are used: Word wrap with centering is selected by default. This was done for the Omnia 9 that uses the Stereo Tool RDS encoder the behavior is now identical for both systems. RDS: Ignoring trailing spaces for word wrap. Added a lot of presets. Added interface for blind users. Set environment variable STBLIND to 1 or YES. Reduced latency: Latency: ASIO: Add processing priority control to enable very low sound card latency ( 1-3 ms) without hiccups total latency of 16 ms possible. Latency: Upsampling (for FM) and downsampling (for high input sampling frequencies) no longer causes extra latency. Latency: Reduced Composite Limiter latency from 1.7 ms to 0.9 ms at latencies 512 and 1024. Some (very small) effect on audio. Latency: Found a bug that caused hiccups in audio processing, fixed it. Results in lower possible latency. Latency: Pushing data to the output before the processing instead of afterwards to reduce latency. Improved audio quality at low latencies: Dynamically adjusting audio processing windows at low latencies to better match the type of audio that comes in. Dynamically adjusting clipping effectiveness to avoid clipping too much in certain conditions, depending on audio processing window and audio content. This protects soft highs against horrible vibration caused by bass sounds at latency 512 and (less) 1024. Upsampling and downsampling: Far less artifacts caused, far cleaner output signal. Created alternative Phase Rotation filter for low latencies (512, 1024). Multiband: Added an artifact protection step that limits the maximum difference in reduction between adjacent bands, if they cause too much (configurable) artifacts. Multiband: Adapted frequency bands and band content to make Multiband at latency 512 sound as much as possible as higher latencies. Loudness: Disabled Improved Bass Distortion Protection filter for latency 512 because it caused distortion. Loudness: Fixed vibrations caused by Very deep bass distortion protection at latency 512 and 1024. Now also enabled for latency 512. Hard Limit: Fixed (minor) artifacts in upsampled audio. Bass Boost: Reduced artifacts at latency 512. Note: Using a less steep filter (bigger difference between frequencies) helps Other changes: Loudness: Improved Punch filter: Now causes far less artifacts than before. But it is less effective for soft sounds. Loudness: Removed distortion caused by Punch. Bass Boost: Removed distortion. Quality and Latency are now part of presets, and STS files. Bug fix: Loading and saving Noise levels in VST version resulted in incorrect behavior. Bug fix: Displayed Multiband output levels were unreliable at low latencies, especially clipping. Bug fix: For high input sample rates, now removing very high frequencies that cause issues with Hard Limit. ASIO: Added ASIO Configure button, which allows setting ASIO parameters such as ASIO buffer granularity. ASIO: Changed buffer size configuration to match whole ASIO grains. Added and updated a number of presets. Noise gate: Fixed crackling sounds in first band. Added FM Hiss filter. Removes FM hiss from input signal, useful for re-transmitting FM signals. AGC: Added Based on volume before Pre Amp checkbox to make it easier to create presets with consistent behavior at different Pre Amp settings. AGC bug fix: Behavior was different at different input sample rates. AGC: Lock band 1 to band 2 if the bass is loud but not extremely loud it is not lowered. This avoids sound changes caused by the AGC, and results in a warmer, more natural sound. Multiband: Added Flat frequency response mode which works much better. Multiband: Added display of the median amplification, useful for configuring Multiband settings. Singleband: Added AGC style mode. Doesnt seem too useful though. Final Limiter: DIFF mode fixed. Final limiter is now automatically bypassed when Loudness is used (improves audio quality and reduces CPU usage). Loudness: Added filters to protect highs against loud bass sounds. The bass level can now be set much higher without causing distortion, even if Punch is used. Loudness: Added maximum mirror reflection to make the sound more natural (less metalic) and louder. Loudness: Fixed Improved loudness distortion protection artifacts. Loudness: Fixed Very deep bass distortion protection artifacts. Loudness: Removed Allow louder lows slider (no longer needed). Loudness: De-esser: Replaced Allow louder highs by an AGC. Loudness: Added a slider to increase the dynamics of the output (by allowing some more harmonic distortion). Loudness: Added a filter to protect certain sounds (female voices) against vibrating. Loudness: Fixed dynamic Loudness input level behavior of Punch Hard Limit: Added an extra slower Hard Limit phase volume comes up slower when the peaks are really loud. Added Lossy Compression (MP3AACOGG. ) output optimization filter for webradio stations. See this image for a display of the effect of different pre-emphasis values on 128 kbits MP3 files this image displays a frequency analysis (high frequency content means clipping), again 128 kbits MP3, Post Amp at 90. FM Output: Repaired the lowpass filter which sometimes caused artifacts. VST plugin notifies host of the delay. VST plugin: Bypass mode fixed. VST plugin: Removed separate window border, now loading inside host window. VST plugin now also works in mAirList. Improved handling of hiccups in the stand alone version when the CPU load gets very high. Presets: Made Generic and Web Radio presets sound much closer to the sound of modern CDs. Presets: Made FM presets sound much closer to that of popular hardware processors (more bass and highs, among others). Presets: Added new user presets, removed some obsolete ones. Command line version: Files larger than 2 GB can be processed now. Linux version added Both a standard command line version and one with a GUI. Cleaned up unused code: Reduced executable size by around 40, installer by 25. Reduced CPU load by 3-4. Reduced memory usage by over 4 MB (from over 24 MB to under 20 MB). Removed part of Core Duo support from the SSE2 version (no effect on performance, only increased file size). Removed SSE2 support from the SSE version (which was accidentally included in it). Fixed Windows Vista This program might not have installed correctly warning. VISIT SITE button was being redrawn continuously - fixed. SSE (Pentium 3) version: Replaced some generic code by SSE code (70 performance boost). Multiband: Very large differences between equalizer or soft limit settings in the lowest bands are handled correctly now. AGC: Improved handling of sudden spikes. AGC: Remove remaining peaks gives far less artifacts, sounds more stable. AGC: Introducted dynamic hold time to improve control of output level (much closer for tracks with very different dynamics contents). AGC: Fixed bug that could cause strange behavior durig absolute silence. AGC: Added possibility to work on 2 channels combined, instead of just operating on them separately. AGC: Added gating. AGC: Added 3rd band for improved highs control. Noise gate: Disabled 1st band because it caused artifacts. Completely redesigned the processing pipeline to reduce the maximum latency from over 1.5 seconds to at most 0.1 second, with a better sound quality. Made latency configurable in 4 steps: 512 (ca. 13 ms), 1024 (25 ms), 2048 (47 ms) or 4096 (93 ms, best quality) samples. Reduced CPU usage by more than 50. Replaced 3 quality modes (LO, MID, HI) by a continuous slider. Reduced memory usage by 17 MB. Replaced Pre Limiter by an RMS-based 1 or 2-band AGC, which controls the volume much better. Added a Bass Boost filter. Added Phase Rotation filter that greatly improves the sound quality of many difficult sounds (trumpets, female voices) when loud output levels are required. Added a filter that removes any pre-ringing (weird sounds before a sound burst) which was present in older Stereo Tool versions. Improved the Multiband clippers audio quality: Multiband clipping sound much cleaner now. Removed Trumpet sound filter from earlier versions that was intended to improve the quality of certain sounds, but also caused a lot of side effects (no longer needed due to the new phase rotation filter). Removed Multiband HQ mode (no longer needed). Made main window resizable and maximizable. Bug: No 2 Stereo Tool instances can be loaded in a VST engine based using the same Stereo Tool DLL file. A popup now displays this message and gives a workaround. Presets updated (now using Bass Boost and Phase Rotation where appropriate) or added. RDS without stereo no longer generates a 19 kHz pilot tone. In case of FM output, normal output is closer to the FM sound, and the FM sound is clipped better before FM Overdrive - using FM Overdrive now leads to a 3 louder output signal than before with better quality. Input and output volume meters blink in red to indicate clipping if the maximum volume is reached (input) or surpassed (output). Improved performance of waveform and MPX display. Improved dithering (final step in processing): If audio is not changed, dithering doesnt change the sound anymore. Bug fix: VST version now allows opening multiple instances of Stereo Tool (earlier versions crashed). Bug fix: Hangups when playing. WAV or lossless files in Winamp. Bug fix: RDS AF (Alternative Frequencies) was not turned off properly if no frequencies are specified. It might have worked, or no, depending on the receiver. Added warnings about sound quality when latency is reduced or FM pre-emphasis is enabled. Configure Audio and Display button blinks when audio is being removed due to the Clear buffer if no data arrives for setting. Waveform display indicates Bypass mode. Improved looks of tray icon. Bug fix: Bypass all in ASIO mode (stand alone) now works properly. RDS volume adjusted: It was slightly louder than the setting. Loudness: Added yet another filter (Very deep bass distortion protection) to reduce bass distortion and make the total sound cleaner. Loudness: Now bypassing Punch for frequencies below 45 Hz for cleaner output sound. Loudness: Improved handling of female voices and xylophone-on-top-of-low-strings sounds, leading also to an overall more natural sound. Re-organized Loudness window. Should be clearer now. Stereo Boost: Added a filter that removes excessive reverb that occurs in some songs. The result is much closer to the original sound, but still with strongly increased stereo. Bug fix: VST plugin version: Some version 4.00 parameters were not saved. Slightly improved some presets. Loudness: Added a filter to reduce bass distortion. Bass levels can now be set much higher without getting more distortion than before, or kept at the same level for better sound quality. Loudness: Added a deep bass boost filter that boost very low (up to about 70 Hz) bass sounds by about 1.5 dB (at 100 Hz the bass is about 0.8 dB softer though). Loudness: Added a limiter to the CleanPunch filter to remove distortion caused when setting it close to Punch. Moved CleanPunch default setting closer to Punch. Low Quality mode: Changed the processing that is used for low quality mode. Quality is much better than before, and the CPU load is reduced. Low Latency mode: Fixed some issues in the low latency processing. Latency is now approx. 100 ms lower than before, quality is much closer to that of normal latency processing. Noise gate: Noise reduction can now be set for each frequency band. Added a number of presets with more bass. Added Marinos Urban preset to Web Radio presets. Added new stereo widening filter: Stereo Boost. Updated most of the presets to use the new Stereo Boost filter. Reduced the minimum value for the gap detection time. Bug fix: No audio problem unless bypass is used solved. Bug fix: Too long delay in tray icon volume display solved. Improved Hard Compression: Dance presets. Bug fix: Under certain circumstances occasional hangups could occur. That should be solved now. Single core systems were not properly recognized and used code that was optimized for multicore systems. The CPU load on single core systems should be slightly lower now. Loudness filter: Improved the filter that detects high frequency distortion caused by very low frequencies. Low frequencies are now much louder, and distortion is greatly reduced. Loudness filter: Added a slider to choose between clean sound (with ringing artifacts) or punchy, as-close-as-possible to the original sound. Punchy sounds a lot better, but it may introduce some distortion (which is largely solved by the fix on the previous line). When opening a window that was already open, it jumps to the front now. Stereo Tool logo added to the window title bar, also displayed now when using ALT-TAB. Bug fix: Version 4.01 crashed on some Vista systems. Loudness filter: Improved bass sound, especially for high (3.0) Loudness values. Bug fix: VST plugins working again. Interface: When clicking on a panel which window is already open, it jumps to the front. Redesigned the interface. The main window is now smaller and less crowded. Each filter has its own window now, and many values are displayed in dB also. The output displays in the main window are now synchronized to each other, and can be synchronized to the audio player sound card output. Added a fix for the annoying part of the previous song gets played at the start of the next song problem in Winamp. Note that this fix is a workaround for what is essentially a problem of Winamp, and it may not be perfect. Reorganized the Presets pulldown menu, and added the 10 last used preset files to the list. Also the currently selected preset is visible in the Presets pulldown button now. Added more control over how the Loudness filter works. Its now possible to allow more highs or more lows to pass through (this may cause some distortion though). Pre Limiter is now using perceptive volume instead of peak level for much better initial volume control. Added a filter to remove remaining loud peaks after the Pre Limiter filter. The Pre Limiter works with a delay, the new filter is a lookahead filter that removes remaining peaks, which occur mostly at the start of a new track after some silence. Final limiter now has an option to use less CPU power (in that case it doesnt preserve the volume). Added a pre-final limiter filter for loud peaks (the normal Final Limiter is loudness preserving, but it doesnt handle very loud sounds well. This pre-final limiter filter removes those very loud sounds.) Loudness filter: The Loudness filter can now handle much louder peaks than before. This even makes it possible to use the Loudness filter as a limiter. Loudness filter: Loud S sounds now cause far less bass level reduction - so the sound is more stable. Loudness filter: Added a filter that detects and removes high frequency distortion caused by loud deep bass sounds. Loudness filter: Improved the filtering so it sounds smoother now. And with the same settings the sound is slightly louder than before with less artifacts. RDS: Added Now playing support for Winamp plugins without the need to use external files. RDS: Added date and time displays. Improved the HARD LIMIT filter, sound should more consistent (less pumping) now. Included the Singleband Compressor from versions 2.11 and below again. Its available from the Limiting Loudness window. Added some presets made by Vamprecords, SuperH and Bojcha. Added a Bypass all button on the main window. Added FM RDS TA button on the main window so traffic announcements can now be indicated without opening the FM configuration window. Performance: On multicore systems, the processing now uses multiple threads. Some slower AMD dualcore PCs couldnt handle FM output in High quality, now they can. Fixed a bug: The highpass filter seems to have been disabled when Multiband compression is turned on since version 3.50. Fixed a bug that could cause blocks of bad pixels in the user interface. Fixed a bug that caused presets to sound slightly different when they were selected while audio was playing. Fixed a bug: On startup, even if Multiband HQ was ticked Multiband didnt run in HQ mode. Greatly reduced the gaps that occur when switching between presets. Uncoupled screen redraws from processing. This allows Windows to select a different CPU for screen updates on multicore systems, and they might get disabled temporarily on single CPU systems when the load is high. Loudness filter: Louder and more stable bass sound. Soundcard support (stand alone version, Direct Soundcard Output on plugin versions): No more stuttering when the output buffer is empty, instead a short break is inserted such that the buffer is filled again when playback resumes. (No change for ASIO output). Loudness filter: Warmer, louder and more stable sound with deeper bass and cleaner highs. Added wave input and output displays. Added MPX spectrogram display for FM output. Fixed a bug that caused the Loudness filter in Normal quality mode to sound softer ( 0.3 dB for Loudness 4.0). FM transmitter bug fix: In version 3.52 the improved filtering caused RDS reception problems on some (mainly German) FM receivers. Fixed. Stereo widener is now almost as strong as in versions before 3.51, but with far less artifacts than before. Especially low quality MP3s should sound a lot better. FM transmitters: MPX filtering cleaned up further, noise level is now around 90 dB below the actual signal (used to be about 30 dB). Very minor change in pre-filtering steps that should slightly improve the handling of loud high frequencies. FM transmitters: Improved filtering for a cleaner MPX signal. Stereo widener causes less artifacts, but also has a bit less effect. Loudness filter: Recalibrated to achieve a much cleaner and louder sound. Distortion that was still present in version 3.40 is now nearly gone, and values up to 3.00 sound good. Values up to 4.00 are now allowed. Loud highs now have far less effect on the loudness of the other sounds - so it sounds much more stable. Added a filter to improve trumpet sounds. Improved quality and loudness of deep bass sounds. Improved FM pre-emphasis to match the official specification (in older versions the pre-emphasis wasnt strong enough, so this new version sounds brighter.) Volume slider now show both pre-emphasized and de-emphasized output levels. Changed Multiband compressor to get a warmer output sound with lower peak levels. Improved the HARD LIMIT filter to sound cleaner, cause far less pumping, and to achieve a louder output volume. Hard Limit filter is now also used if Direct Sound Card Output is used. Improved the composite limiter filter to cause far less pumping, and to achieve a louder and cleaner output volume. Updated presets, added FM Extreme preset. Improved Lowpass filter to filter closer to the selected lowpass frequency (important for AM presets). Also the filtering quality is now much better - the spectrum above the filtering frequency is now really empty. Bug fix: Fake stereo didnt work in version 3.40 - it does work again now. The CPU load is about 5-10 higher than that of version 3.40 - measured using the FM Loud preset. Loudness filter: Completely rewritten. Now reaches much louder sounds (up to 12 dB) with far less effects on the sound quality. Even notoriously difficult sounds such as xylophones are now handled without introducing distortion. Added FM transmitter calibration. If you are having problems with your FM signal (caused by low quality transmitter equipment, sound card, long cables etc.), you can now configure Stereo Tool to accomodate for that. (On my system, stereo separation is increased by 15 dB after calibrating) Composite limiter now operates at 705.6 or 768 kHz. This removes peaks that were introduced by the lowpass filter of the sound card. Hard limit now operates at 352.8 or 384 kHz. This removes peaks that were introduced by the lowpass filter of the sound card. Removed the Loudness option because it is no longer needed. Tweaked the presets to take advantage of the improved Loudness filter (more punch). The CPU load is roughly identical to that of version 3.30 in Low and Normal quality mode as long as the composite limiter oversampling is not used. And it will be a lot slower on PCs without SSE2 support. Composite limiter added. Gives about 7 (0.5 dB) extra loudness on FM transmitters, without affecting the FM Pilot and RDS signals. . Memory footprint reduced by about 12 MB (3.29 used 34 MB, 3.30 uses 22 MB)
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